Transceiver method and signal therefor embodied in a carrier wave for a frame-based communications network

ABSTRACT

A method and signal therfor embodied in a carrier wave for sending information from transmit stations to receive stations over a transmission medium of a frame-based communications network. The information is sent in transmit frames having a frame format comprising a fixed rate header, followed by a variable rate payload, followed by a fixed rate trailer. The fixed rate header includes a preamble. The preamble has a repetition of four symbol sequences for facilitating power estimation, gain control, baud frequency offset estimation, equalizer training, carrier sensing and collision detection. The preamble also includes a frame control field. The frame control field has scrambler control information for frame scrambling initialization, a priority field to determine the absolute priority a transmit frame will have when determining access to the transmission medium, a payload encoding field which determines constellation encoding of payload bits in the variable rate payload, and a header check sequence for providing a cyclic redundancy check. The variable rate payload is transmitted pursuant to dynamic adjustable frame encoding parameters for improving transmission performance for a transmit frame being transmitted from a transmitting station to a receiving station. The header also includes a destination address field, a source address field and an ethertype field.

CROSS-REFERENCE TO RELATED APPLICATION(S)

This patent application claims the benefit of the filing date of U.S.Provisional Patent Application No. 60/197,224 filed Apr. 14, 2000; andU.S. Provisional Patent Application No. 60/196,002 filed Apr. 7, 2000;the entire contents of both of which are hereby expressly incorporatedby reference.

This patent application is further related to the following U.S. patentapplications filed concurrently herewith and commonly assigned, entitled“A Method of Sharing Information among a Plurality of Stations in aFrame-based Communications Network”, application Ser. No. 09/825,708, “AMethod of Enhancing Network Transmission on a Priority-enabledFrame-based Communications Network”, application Ser. No. 09/825,897, “AMethod of Determining a Start of a Transmitted Frame in a Frame-basedCommunications Network”, application Ser. No. 09/825,903, “A Method ofDetermining an End of a Transmitted Frame in a Frame-basedCommunications Network”, application Ser. No. 09/825,775, “A Method forProviding Dynamic Adjustment of Frame Encoding Parameters in aFrame-based Communications Network”, application Ser. No. 09/826,218, “AMethod for Selecting Frame Encoding Parameters in a Frame-basedCommunications Network”, application Ser. No. 09/826,435, “A Method forSelecting Frame Encoding Parameters to Improve Transmission Performancein a Frame-based Communications Network”, application Ser. No.09/825,756, “A Method of Determining a Collision Between a Plurality ofTransmitting Stations in a Frame-based Communications Network”,application Ser. No. 09/825,801, “A Method of Providing SynchronousTransport of Packets Between Asynchronous Network Nodes in a Frame-basedCommunications Network”, application Ser. No. 09/825,851, “A Method ofControlling Data Sampling Clocking of Asynchronous Network Nodes in aFrame-based Communications Network”, application Ser. No. 09/826,067, “AMethod for Distributing Sets of Collision Resolution Parameters in aFrame-based Communications Network”, application Ser. No. 09/825,689, “AMethod and Apparatus for Optimizing Signal Transformation in aFrame-based Communications Network”, application Ser. No. 09/825,599, “AMethod and Apparatus for Transceiver Noise Reduction in a Frame-basedCommunications Network”, application Ser. No. 09/825,638, and “A Methodfor Selecting an Operating Mode for a Frame-based CommunicationsNetwork”, application Ser. No. 09/825,791.

BACKGROUND OF THE INVENTION

The present invention relates to the field of communications, and, inparticular, to a frame-based communications network.

As computers become more and more cost effective for the everydayconsumer and for small businesses, such computers become more plentifulfor use within local area environments such as homes, office buildingsand the like. For example, within a home a person with a computer in thebedroom, and another in the living room, may want to share common files,utilize a common digital subscriber line (DSL), or otherwise transferinformation between the computers. Accordingly, various technologies arebeing developed for computer interconnection of multiple computerslocated within such environments. One example of such technologies arethe Home Phoneline Network Alliance (HPNA) specifications for local areanetwork (LAN) computer interconnection which utilize existing telephonelines within the local environment for the transmission of data packetsbetween the computers.

FIG. 1 a shows in block diagram form a general home networkingenvironment within which the present invention can be implemented. Homenetwork 10 includes existing (installed) plain old telephone service(POTS) wiring 12, network clients 14, the computer port side of modem 16and fax 18. POTS wiring 12 provides wiring infrastructure used tonetwork multiple clients at a customer premises (e.g., home) 20. POTSwiring 12 can be conventional unshielded twisted pair (UTP) wiring thatis generally routed internally in the walls of the customer premises 20to various locations (e.g., rooms) within the customer premises.Subscriber loop 22 (also called a “local loop”) is a physical wiringlink that directly connects an individual customer premises 20 to theCentral Office through telephone network interface 24, a demarcationpoint between the inside and outside of customer premises 20. Ofparticular importance for residential networks are systems that providecommunication between computers as reliably and with as high a data rateas possible. Communication over residential telephone wiring is providedthrough inventive frame-oriented link, media access and physical layerprotocols and implementation techniques associated therewith describedherein.

Referring now to FIG. 1 b, those skilled in the art can appreciate thathome phone-line network configuration 10 can also utilize interface 6010to provide signals outside customer premises 20. For example, interface6010 can include a V.90 modem as described above, connected through thecentral office to an internet service provider. Interface 6010 caninclude an ADSL modem, a VDSL modem or the like transport interface.

Another desired solution for high speed data communications appears tobe cable modem systems. Cable modems are capable of providing data ratesas high as 56 Mbps, and is thus suitable for high speed file transfer.In a cable modem system, a headend or cable modem termination system(CMTS) is typically located at a cable company facility and functions asa modem which services a large number subscribers. Each subscriber has acable modem (CM). Thus, the CMTS facilitates bidirectional communicationwith any desired one of the plurality of CMs. Referring to FIG. 1 c, ahybrid fiber coaxial (HFC) network 1010 facilitates the transmission ofdata between a headend 1012, which includes at least one CMTS, and aplurality of homes 1014,. each of which contains a CM. Such HFC networksare commonly utilized by cable providers to provide Internet access,cable television, pay-per-view and the like to subscribers.Approximately 500 homes 1014 are in electrical communication with eachnode 1016, 1034 of the HFC network 1010, typically via coaxial cable1029, 1030, 1031. Amplifiers 1015 facilitate the electrical connectionof the more distant homes 1014 to the nodes 1016, 1034 by boosting theelectrical signals so as to desirably enhance the signal-to-noise ratioof such communications and by then transmitting the electrical signalsover coaxial conductors 1030, 1031. Coaxial conductors 1029 electricallyinterconnect the homes 1014 with the coaxial conductors 1030, 1031,which extend between amplifiers 1015 and nodes 1016, 1034. Each node1016, 1034 is electrically connected to a hub 1022, 1024, typically viaan optical fiber 1028, 1032. The hubs 1022, 1024 are in communicationwith the headend 1012, via optical fiber 1020, 1026. Each hub istypically capable of facilitating communication with approximately20,000 homes 1014. The optical fiber 1020, 1026 extending intermediatethe headend 1012 and each hub 1022, 1024 defines a fiber ring which istypically capable of facilitating communication between approximately100,000 homes 1014 and the headend 1012. The headend 1012 may includevideo servers, satellite receivers, video modulators, telephone switchesand/or Internet routers 1018, as well as the CMTS. The headend 1012communicates via transmission line 1013, which may be a T1 or T2 line,with the Internet, other headends and/or any other desired device(s) ornetwork.

Given the HPNA environment and the Cable Modem System environment, anopportunity exists for a system provider to integrate each respectiveenvironment with voice services. FIG. 1 d depicts such an integratedenvironment. As can be seen in FIG. 1 d, a connection point in the hometo the telephony world (e.g., the world of video, voice, high speed datanetwork traffic), could be provided to a home user through cable modem1046 which would include an HPNA transceiver. The cable modem systemprovider may also wish to accomodate providing telephone service alongwith high speed data service. A home computer user, rather than using atraditional modem to connect to an internet service provider, would findit convenient to utilize cable modem 1046, taking advantage of the veryhigh speed data service provided by the cable modem. Having a cablemodem customer, the cable modem provider may also find it commerciallybeneficial to offer video feeds, and telephone service over the samecable modem network.

A cable modem having an HPNA V2 transceiver included therein, canreadily interface into the home phone line network through the telephonejack within the home. Computers coupled to the home network thencommunicate through the cable modem to the outside telephony world asdescribed above. Telephone service coming from outside the customerpremises over the cable modem system would be in a digitized packetizedformat. It would then proceed over the HPNA network in the samedigitized pocketing format. If the user, in addition to having computersand the like attached to the HPNA network, wished to have an analogtelephone(s) connected to the HPNA, the telephone'(s) analog signalwould go through a digital conversion and put the digital informationinto packets for passing the packets back and forth over the network.The analog telephone signal is sampled and packetized at the appropriateclock rate creating the packet after a certain number of samples.

Therefore, to effectively operate in such communications networkenvironments a need exists for a method for distributing sets ofcollision resolution parameters in a frame-based communications network.The present invention as described and claimed in this applicationprovides a solution to meet such need.

SUMMARY OF THE INVENTION

In accordance with the present invention a signal embodied in a carrierwave is provided for sending information from transmit stations toreceive stations over a transmission medium of a frame-basedcommunications network. The information is sent in transmit frameshaving a frame format comprising a fixed rate header, followed by avariable rate payload, followed by a fixed rate trailer.

Also in accordance with the present invention the fixed rate headerincludes a preamble. The preamble has a repetition of four symbolsequences for facilitating power estimation, gain control, baudfrequency offset estimation, equalizer training, carrier sensing andcollision detection. The preamble also includes a frame control field.The frame control field has scrambler control information for framescrambling initialization, a priority field to determine the absolutepriority a transmit frame will have when determining access to thetransmission medium, a payload encoding field which determinesconstellation encoding of payload bits in the variable rate payload, anda header check sequence for providing a cyclic redundancy check. Thevariable rate payload is transmitted pursuant to dynamic adjustableframe encoding parameters for improving transmission performance for atransmit frame being transmitted from a transmit station to a receivestation. The header also includes a destination address field, a sourceaddress field and an ethertype field.

Further in accordance with the present invention a method fortransmitting a transmitting frame embodied in a carrier wave fromtransmit stations to receive stations over a transmission medium of aframe-based based communications network is provided. One or moretransmit stations are coupled to the transmission medium, each transmitstation transmitting frames having a frame format including a fixed rateheader, followed by a variable rate payload, followed by a fixed ratetrailer. One or more receive stations are coupled to the transmissionmedium, each receive station upon receiving a received framecorresponding to the transmitting frame addressed to the receivestation: (1) detects a start of the received frame utilizing apredefined preamble format for the transmitting frame having a pluralityof identical copies of a preamble symbol sequence transmittedsequentially, (2) decodes the received frame (3) measures and tracksperformance of frame decoding, (4) determines network performancecharacteristics for establishing desired performance based uponmeasuring and tracking the performance of the frame decoding, (5)indicates to the transmit station changes to payload encoding parametersin the fixed rate header based upon determining network performanceimprovement characteristics, wherein the transmit station changes thepayload encoding parameters in the fixed rate header for encoding nextfuture transmitting frames (6) and determines whether a collisionbetween two or more transmit stations occurred at one of the transmitstations utilizing an estimate of error power of defined copies of thepreamble symbol sequence.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1 a, 1 b, 1 c and 1 d are simplified block diagrams showing a homenetworking environment within which the present invention can beimplemented.

FIG. 2 is a seven-layer network stack model, according to the ISOseven-layer network standard, as used in accordance with the presentinvention.

FIGS. 3 a and 3 b show a broadcast/multipoint network and apoint-to-point network, respectively, for use in accordance with thepresent invention.

FIGS. 4 a and 4 b show respectively an integrated MAC/PHY aspect and ananalog front end aspect of an embodiment of the present invention.

FIG. 5 depicts in block diagram form a transmitter aspect of anembodiment of a PHY in accordance with the present invention.

FIG. 6 shows the frame format in accordance with the present invention.

FIG. 7 depicts the frame control field in accordance with the presentinvention.

FIG. 8 shows the frame control fields transmission order in accordancewith the present invention.

FIG. 9 depicts the payload encoding values in accordance with thepresent invention.

FIG. 10 shows fields as described in IEEE Standard 802.3.

FIG. 11 depicts the frame-synchronized scrambler in accordance with thepresent invention.

FIGS. 12 a–12 g show the bit-symbol mapping performed by theconstellation encoder in accordance with the present invention.

FIG. 13 depicts the relative scaling of different constellations at asingle baud rate in accordance with the present invention.

FIG. 14 illustrates 2 MBaud to 4 MBaud and 4 MBaud to 2 MBaudtransitions in accordance with the present invention.

FIG. 15 shows in simplified block diagram form an example QAMimplementation in accordance with the present invention.

FIG. 16 depicts compatibility mode frame formating in accordance withthe present invention.

FIG. 17 shows the transmitter aspect of the PHY embodiment operating incompatibility mode in accordance with the present invention.

FIG. 18 depicts the format of each subframe and gap for the 2-Mbaudheader in compatibility mode in accordance with the present invention.

FIG. 19 shows the format for all of the last of the 4-Mbaud payloadsubframes and gaps in compatibility mode in accordance with the presentinvention.

FIG. 20 shows the EOF/EOP symbol sequence for the 2-Mbaud payload casein compatibility mode in accordance with the present invention.

FIG. 21 shows the EOF/EOP symbol sequence for the 4-Mbaud payload casein compatibility mode in accordance with the present invention.

FIG. 22 depicts the compatibility mode frame in conjunction with atheAccess ID interval in accordance with the present invention.

FIGS. 23 a and 23 b depict the metallic power spectral densityassociated with the transmitter in accordance with the presentinvention.

FIG. 24 shows the magnitude of the transmitter output in accordance withthe present invention.

FIGS. 25 and 26 depict maximum peak-to-peak interferer level overfrequency range in accordance with the present invention.

FIG. 27 shows minimum impedance over frequency range.

FIG. 28 shows an example of input impedance in view of a lower boundmask over frequency range in accordance with the present invention.

FIG. 29 depicts the MAC logical layers and corresponding functions inaccordance with the present invention.

FIG. 30 shows in functional block diagram form an embodiment of atransceiver in accordance with the present invention.

FIG. 31 depicts a valid frame transmission with respect to the carriersense function.

FIGS. 32 and 33 depict signal and priority slots involved withtransmission and collision aspects in accordance with the presentinvention.

FIGS. 34 a and 34 b show transmission aspects without and with priorityaccess.

FIG. 35 shows length of collisions and non-collisions.

FIG. 36 indicates various MAC parameters in accordance with the presentinvention.

FIGS. 37 and 38 depicts basic formats for link control frames, longsub-type and short sub-type, respectively, in accordance with thepresent invention.

FIG. 39 shows rate request control frames in accordance with the presentinvention.

FIG. 40 indicates the assigned values that may appear in the banddescription entries in the rate request control frames in accordancewith the present invention.

FIG. 41 indicates the values that may appear n the OpCode entry in therate request control frame in accordance with the present invention.

FIG. 42 indicates further rate request control frame terms anddefinitions.

FIGS. 43 a and 43 b show state diagrams and table involving linkintegrity functionality in accordance with the present invention.

FIG. 44 depicts a link integrity short frame in accordance with thepresent invention.

FIG. 45 indicates a compatibilites and status announcements controlframe in accordance with the present invention.

FIG. 46 shows compatibilities and status announcements flag definitionsin accordance with the present invention.

FIGS. 47 and 48 depict variable and timers, respectively, in accordancewith the capabilities and announcements functionality.

FIG. 49 indicates basic sets of status and priority information inaccordance with the capabilities and announcements functionality.

FIG. 50 shows composite sets in accordance with the capabilities andannouncements functionality.

FIGS. 51 a and 51 b show examples link layer priorities in accordancewith the present invention.

FIGS. 52 a–52 f.2 depict various LARQ frame information in accordancewith the present invention.

FIG. 53 shows variables and parameters involved with the LARQ senderoperation in accordance with the present invention.

FIG. 54 shows variables and parameters involved with the LARQ receiveroperation in accordance with the present invention.

FIGS. 55 a and 55 b depict vendor specific formats with regard to linklayer protocol in accordance with the present invention.

FIG. 56 shows state information for carrier sense decision logic inaccordance with the present invention.

FIG. 57 depicts an embodiment of the low-delay detector of the carriersensor in accordance with the present invention.

FIG. 58 shows an example of an averaging circuit for L*n=16 samples ofthe carrier sensor in accordance with the present invention.

FIG. 59 depicts the robust detector of the carrier sensor in accordancewith the present invention.

FIG. 60 shows the first test of the end-of-carrier detector of thecarrier sensor in accordance with the present invention.

FIG. 61 depicts a state diagram involving an embodiment of the carriersensor.

FIGS. 62 a and 62 b show course dB table values and fine dB tablevalues, respectively, of an example embodiment in accordance with thepresent invention.

FIGS. 63 a, 63 b and 63 c show a preamble, channel and characterizationsignal in accordance with the present invention.

FIG. 64 graphically depicts a template signal in accordance with thepresent invention.

FIG. 65 shows a comparison circuit to provide match/no match assessmentin accordance with the present invention.

FIGS. 66 a and 66 b show respectively unsigned 7.2 values and unsigned3.2 values in accordance with the present invention.

FIG. 67 shows an embodiment of a power estimation subcircuit inaccordance with the present invention.

FIG. 68 shows a MMSE FSE/DFE structure in accordance with the presentinvention.

FIG. 69 depicts a split winding transformer in accordance with thepresent invention.

FIG. 70 shows a VoIP system in accordance with the present invention.

FIG. 71 shows packet arrival timing relationships in accordance with thepresent invention.

FIGS. 72 a and 72 b show transmit queues before and after priority framereordering respectively in accordance with the present invention.

FIG. 73 depicts a VoIP system in accordance with the present invention.

FIGS. 74 and 75 show upstream and downstream latency components inaccordance with the present invention.

FIG. 76 shows the Timestamp Sync Frame format in accordance with thepresent invention.

FIG. 77(1)-77(2) show the Timestamp Report Frame format in accordancewith the present invention.

FIGS. 78 and 79 show MAC pin functionality in accordance with thepresent invention.

FIG. 80 shows a timing recovery circuit in accordance with the presentinvention.

FIG. 81 depicts DPLL jitter in accordance with the present invention.

FIG. 82 shows a limited timing recovery circuit embodiment in accordancewith the present invention.

FIGS. 83 a–83 g depict various pin and bit locations in accordance withthe present invention.

FIGS. 84 a and 84 b show DPLL output jitter in accordance with thepresent invention.

FIG. 85(1)-85(3) shows a Timestamp Report Message in accordance with thepresent invention.

FIG. 86 shows a Timestamp Request Message in accordance with the presentinvention.

FIG. 87 shows a Timestamp Slot Request Message in accordance with thepresent invention.

FIG. 88 depicts the ITU G.712 specification for total distortion inaccordance with the present invention.

FIGS. 89 a, 89 b and 89 c show various SNR in accordance with thepresent invention.

FIG. 90 shows jitter clock characteristics in accordance with thepresent invention.

FIGS. 91 and 92 show, respectively, ADC and DSC data paths of an analogtest chip in accordance with the present invention.

FIG. 93(1)-93(2) show various CSA flags in accordance with the presentinvention.

FIG. 94 shows the form for CSA extension for CSS in accordance with thepresent invention.

FIG. 95 depicts desired codings for the CSS register bits in accordancewith the present invention.

FIG. 96 depicts a PCOM field utilized in accordance with mode selectionaspects of the present invention.

FIG. 97 depicts the relative precedence of variable employed in modedetermination in accordance with the present invention.

DETAILED DESCRIPTION OF THE INVENTION

Before addressing the Voice aspects of the present invention, theHomenetworking implementation aspects will be first addressed.

Homenetworking Implementation Aspects

A communications network typically includes a group of nodesinterconnected by a transmission medium. The term “node” relates to anydevice that shares frames of data with other nodes in the network.Devices that may make up a node are computers, printers, scanners, etc.A node may also be a telephone, a television, a set-top box fortelevisions, a camera or other electronic sensing or communicationdevice. Any device that can send and/or receive frames of data withother devices via a communication medium may be a node for purposes ofthe present invention.

The transmission medium that links each node in a network is equally oneof a diverse family of media. Common media used include unshieldedtwisted pair (e.g. phone wire, CAT-5 cabling), power lines, opticalfiber, coaxial cable and wireless transmission media. The operationsthat each individual node performs in order to access data from, andtransmit data to, the rest of the network may be logically broken downinto seven layers according to the ISO Open Systems Interconnection(OSI) seven-layer network model, which is also referred to as the“network stack”. The seven layers, from the bottom to the top are: 1)the PHYSICAL layer, 2) the DATA LINK layer, 3) the NETWORK layer, 4) theTRANSPORT layer, 5) the SESSION layer, 6) the PRESENTATION layer, and 7)the APPLICATION layer. FIG. 2 illustrates the ISO seven-layer referencemodel.

The PHYSICAL layer, or physical link layer, or PHY, is concerned withtransmission of unstructured bit stream traffic over physical media, andrelates to the mechanical, electrical, functional, and proceduralcharacteristics to access and receive data from the physical medium. TheDATA layer, sometimes referred to as the data link layer, provides forthe reliable transfer of information across the physical link. It isconcerned with sending frames, or blocks of data, with the necessarysynchronization, error control, and flow control. The NETWORK layerseparates the uppermost layers from the transmission and switchingtechnologies used to connect nodes. It relates to establishing,maintaining, or terminating connection between nodes.

The TRANSPORT layer relates to reliability and transparency in datatransfers between nodes, and provides end-to-end error recovery and flowcontrol. The SESSION layer provides control to communications betweenapplications, and establishes, manages, and terminates connectionsbetween cooperating applications. The PRESENTATION layer providesindependence to the application processes from differences in datasyntax or protocols. Finally, the highest layer, the APPLICATION layer,provides access to the OSI environment for users. Much more has beenwritten about the benefits and distributed functionality of such anarrangement of layers and need not be recounted here.

In frame-based networks, there are two fundamental models ortopologies: 1) broadcast/multipoint networks, where all nodes arephysically attached to the same network medium, and use a single, sharedchannel and frames transmitted on the network are visible to all nodes;and 2) point-to-point networks, where pairs of nodes are connected toeach other with communication channels which are not connected to anyother nodes on the network. Frames transmitted on one channel are notvisible to nodes on other channels unless the frames are retransmittedonto the other channels by a node that is connected to multiplechannels. Each channel may use a separate segment of the network medium,or multiple channels may share a single segment using e.g., FrequencyDivision Multiplexing or Time Division Multiplexing techniques. Onecommon example of such a point-to-point network topology is that usedfor IEEE 10BaseT 802.3 networks, with network nodes connected viapoint-to-point Category 5 unshielded twisted pair cable, usingmulti-port devices called hubs to retransmit frames received from onenetwork segment to all other segments.

FIGS. 3 a and 3 b show a broadcast/multipoint network and apoint-to-point network, respectively, for use with the presentinvention. In FIG. 3 a, representative nodes 140 a, 140 b, 140 c arecommunicatively coupled with a common transmission medium 250 throughindividual segments 240 a, 240 b, 240 c respectively. Thus, a messagecontaining a broadcast destination address sent from one node is sent toall other nodes coupled with transmission medium 250. In FIG. 3 b, nodes140 d, 140 e, 140 f are communicatively coupled to each other byindividual segments 260 d, 260 e, 260 f respectively of transmissionmedia and hub 255. Messages sent from one node to another node on onesegment are not visible to nodes on other segments unless they areretransmitted by a node that is connected to multiple segments, such ashub 255 in a network. Segments 240 a, 240 b, 240 c and commontransmission medium 250 may be (but are not restricted to) a phone line,a power line, a wireless medium, coaxial cable, or a fiber optic medium.Reference to FIGS. 3 a and 3 b should be made with respect to thedescription of the embodiments of the invention as set forth below.

Each node in either type of network has within it a device that permitsthe node to send and receive data frames in the form of electrical,electromagnetic, or optical signals. The device is conventionally asemiconductor device implementing the PHYSICAL layer of the networkconnectivity, and the medium access control (MAC) portion of the DATAlayer of network connectivity.

Returning to FIG. 2, there is shown a basic network illustrating anetwork communication protocol between first node 102 that runs anapplication (“APP X”) and another node 104 that runs the same ordifferent application (“APP Y”). Nodes 102 and 104 communicate message108 via transmission medium 106. In the example shown in FIG. 2, whennode 102 has message 108 to send to node 104, it transfers the messagedown through its network stack on the left, from layer to layer.Application header (AH) 103 is appended to message 108 in theAPPLICATION layer, to identify the application being executed by node102. Original message 108, plus the application header AH, is passed tothe PRESENTATION layer, where it is again appended with a presentationlayer header (PH) 105. Such process continues, accordingly addingsession header (SH) 107, transport header (TH) 109 and network header(NH) 111 down to the DATA layer, where the message and appended headersis encapsulated with data layer header (DH) 112 start of frame (SOF)indicator 113. The DATA layer also may add data trailer (DT) 114 and endof frame (EOF) indicator 115. Data layer header 112 may include a sourceaddress (SA) to identify node 102 sending the message, and may alsoinclude a destination address (DA) to identify the intended recipient orgroup of recipients.

The message with appended headers, trailers and indicators is thenpassed to the PHYSICAL layer where it is passed on to networktransmission medium 106. When received by node 104, the reverse processoccurs in the network stack of node 104. At each layer, the headerand/or trailer information is stripped off as message 108 ascends thenetwork stack.

The details of the network stack in FIG. 2 are provided for referenceonly, and the present invention is not limited to functioning withnetwork stack implementations that exactly match FIG. 2.

Referring still to FIG. 2, the lower two layers are described in furtherdetail. It should be understood that these layers are typicallyimplemented as a combination of logic and memory storage that isconfigured to carry out the task of the layer. The logic can be in theform of hardware, software, firmware, or a combination of those. Eachlayer may also be implemented using programmable gate array (PGA)technology, such as system programmable gate arrays (SPGA) and fieldprogrammable gate arrays (FPGA). Also, each layer, or a combination ofthe layers, may be implemented as an integrated circuit or softwareprogram. Therefore, it should be apparent to those skilled in the art,that there are many ways in which to implement the inventions describedherein.

FIG. 2 shows DATA layers 120 a, 120 b and PHYSICAL layers 220 a, 220 bfor a representative pair of nodes 140 a, 140 b according to theinvention. Each node has within it semiconductor device(s) thatimplement the PHYSICAL layer as well as the medium access control (MAC)and Link Layer portions of the DATA layer, such as that implemented bythe Broadcom Corporation Model BCM 4210 Controller. As discussed above,the PHYSICAL layer is concerned with transmission and reception of bitstream traffic to and from the transmission medium. Transmitters andreceivers, described in more detail below, form a transmission mediuminterface, and may be implemented as a single device or separatedevices.

Referring now to FIGS. 4 a and 4 b, an embodiment implementing theinventive concepts is depicted wherein, for example, a device such ascomputer 14 can be interconnected therethough to premises UTP wiring asset forth in FIG. 1 a, and through which the protocol set forth in FIG.2 is processed. FIG. 4 a shows in block diagram form the controlleraspects of the embodiment, while FIG. 4 b show typical network interfacedevice (NID) analog front end aspects of the embodiment.

Referring to FIG. 4 a, controller 300 is a fully integrated MAC/PHYdevice that transmits and receives data (e.g., 10 Mbps and above asimplemented by the aforementioned Broadcom Corporation Model BCM 4210,4211, 4413 controllers). Controller 300 includes bus interface 310, suchas a PCI or MSI bus interface for communication in accordance withwell-known PC—based and/or peripheral/internet appliance architectures.Controller 300 also includes digital PHY 320 having a FDQAM/QAMtransmitter and receiver interfacing with the analog front end and MAC330, coupling to bus interface 310 through transmit (TX) FIFO 340 andreceive (RX) FIFO 350. Bus interface 310 also has the capability ofsimilarly communicating with other devices 360, such as a v.90 modemthrough v.90 modem interface or a 10/100 Fast Ethernet bus through a10/100 Fast Ethernet interface, and their respective transmit (TX) FIFO370 and receive (RX) FIFO 380. The operations of such bus interfaces andTX/RX FIFOs are well known in the art and are not described in moredetail. The operation of the MAC/PHY aspects of the embodiment aredescribed in more detail herein below.

Referring to FIG. 4 b, NID analog front end 400 connects controller 300depicted in FIG. 4 a to a transmission medium 106 such as a premises UTPwiring as depicted in FIGS. 1 a, 1 b and 1 c. Analog front end 400includes digital input/output (I/O) circuit 410 for transferring samplesand is coupled to a transmit path and a receive path. Digital I/O 410includes clock 412 for driving controller 300 with a 64 MHz +/−100 ppmclock generated by 64 Mhz crystal 414. The transmit path includesdigital-to-analog converter 420 for converting 10 bit sample data to ananalog signal, automatic gain controller 425 for setting gains basedupon input received by digital I/O 410, filter 430, transmit-off switch435, and is coupled to phoneline connector 450, such as a UTP wiringRJ11 connector, through electronic hybrid 440 for buffering signals andfilter/transformer/electronic protection circuit 445. The receive pathincludes analog-to-digital converter 460 for sending valid sample data,variable gain amplifier (VGA) 470, filter 480 for low-passanti-aliasing, VGA 490, and is similarly coupled to phoneline connector450 through electronic hybrid 440 and filter/transformer/electronicprotection circuit 445. Electronic hybrid 440 andfilter/transformer/electronic protection circuit 445 are connectedtherebetween by a plurality of transmit and receive lines (e.g., TX,RX1, RX2) 495. The operations of the analog front end are well known inthe art.

Homenetworking PHY Layer Overview

In accordance with a preferred embodiment of the present invention PHY320 uses 4 MBaud QAM modulation and 2 MBaud Frequency Diverse QAM(FDQAM), with 2 to 8 bits-per-Baud constellation encoding, resulting ina PHY-layer payload modulation rate that ranges from 4 Mb/s to 32 Mb/s.The modulation techniques are set forth in U.S. patent application Ser.No. 09/169,552 entitled “Frequency Diverse Single Carrier Modulation ForRobust Communication Over In-Premises Wiring”, which is incorporatedherein by reference. Information is transmitted on the transmissionmedium/channel in bursts. Each burst or physical layer frame consists ofPHY-layer payload information encapsulated with a PHY preamble, headerand postamble. The PHY-layer payload in each physical frame is that partof the Ethernet Link Level frame that follows the Ethertype fieldthrough the Frame Check Sequence (FCS), plus a CRC-16 and a pad fieldfor the 4 Mbaud rate. Hereafter, “payload” refers to the PHY-layerpayload unless otherwise specified.

Referring to FIG. 5, a transmitter aspect of PHY 320 is shown infunctional block diagram form. Transmitter 500 includes frame processor510, data scrambler 520, bit-to-symbol mapper (constellation encoder)530, and QAM/FDQAM modulator 540. The frame format transmitted bytransmitter 500 is shown in FIG. 6. Frame format 600 consists oflow-rate header section 610, a variable-rate payload section 620, and alow-rate trailer 630. Some parts of the frame are not scrambled, asdescribed below. Except where otherwise stated, all fields are encodedmost significant octet first, least significant bit first within eachoctet. Bit number 0 is the 1sb within a field. Diagrams in the figuresherewith show MSB bits or octets to the left. Header 610 includes apreamble (PREAMBLE64) 612 and is defined as a repetition of four 16symbol sequences (TRN16) that result from encoding 0xfc483084 in theorder described above at 2 MBaud , 2 bits-per-Baud, with the scramblerdisabled. The TRN16 is a white, constant amplitude QPSK sequence. Thepreamble facilitates power estimation and gain control, Baud FrequencyOffset Estimation, Equalizer Training, Carrier Sense, and CollisionDetection as is described in more detail below. Header 610 also includesframe control field 614. Frame control field 614 is a 32-bit fielddefined in the table set forth in FIG. 7. and with the bit-orderingdefined above, the frame control fields are transmitted in the ordershown in FIG. 8. Frame Type (FT) 616 is an eight bit field that isintended to provide flexibility for defining other frame formats infuture versions of the embodiment. Present devices transmit 0 in thisfield, and discard any frames with FT other than zero. All other valuesare reserved. The field definitions in the present embodiment are forFT=0.

The FT field is intended to provide a mechanism for ForwardCompatibility, allowing extensions to use frame formats differing fromthe present embodiment. A next field is scrambler initialization (SI)field 618. A 4-bit field is set to the value used to initializescrambler 520, as described below. A next field is the priority (PRI)field 620 which refers to a media access priority mechanism as describedbelow. The 3 bit PHY priority value (PRI) refers to the absolutepriority that a given frame will be given when determining media access,and is the value used in the MAC embodiment described below. Priority 7has preferential access over Priority 0. PRI field 620 is a fieldcarried in the PHY-level frame transmission and is intended to indicatea 3-bit PHY-level priority or class-of-service indication to thereceiver link level processor for managing priority and class of serviceof the received frame. The PRI value is not used by the receiver PHYprocessor. For stations that do not implement class-of-service the PRIfield is ignored on receive, and is transmitted set to 1. The next field622 is reserved (RSVD)for future use and is ignored by the receiver.Adjacent to field 622 is payload encoding (PE) field 624 whichdetermines the constellation encoding of the payload bits. The PE valuesare defined as forth in FIG. 9. Certain PE values are reserved. ReservedPE values are intended to code for higher Baud rates and carrierfrequencies that will be introduced in later versions of the embodiment.The next field is a header check sequence (HCS) 626. HCS 626 is an 8-bitcyclic redundancy check (CRC) computed as a function of the 128-bitsequence in transmission order starting with the FT bits and ending withthe Ethernet source address (SA) bits, with zeros substituted for theas-of-yet uncomputed HCS field. The encoding is defined by the followinggenerating polynomial.G(x)=x ⁸ +x ⁷ +x ⁶ +x ⁴ +x ²+1Mathematically, the CRC value corresponding to a given frame is definedby the following procedure.

-   -   a) The first 8 bits of the input bit sequence in transmission        order are complemented.    -   b) The 128 bits of the sequence in transmission order are then        considered to be the coefficients of a polynomial M(x) of degree        127. (The first bit of the FT field corresponds to the x¹²⁷ term        and the last bit of the SA field corresponds to the x⁰ term.)    -   c) M(x) is multiplied by x⁸ and divided by G(x), producing a        remainder R(x) of degree <=7.    -   d) R(x) is multiplied by H(x) to produce N(x), where H(x) is        defined as H(x)=x⁷+x⁶+x⁵+x⁴+x²+x+1    -   e) N(x) is divided by G(x), producing a remainder R′ (x) of        degree <=7.    -   f) The coefficients of R′ (x) are considered to be an 8-bit        sequence.    -   g) The bit sequence is complemented and the result is the CRC′.

The 8 bits of the CRC′ are placed in the HCS field so that x⁷ is theleast-significant bit of the octet and x⁰ term is the most-significantbit of the octet. (The bits of the CRC′ are thus transmitted in theorder x⁷, x⁶, . . . x¹, x⁰.) Although HCS 626 is embedded within theprotected bit-stream, it is calculated in such a way that the resulting128-bit stream provides error-detection capabilities identical to thoseof a 120-bit stream with an 8-bit CRC appended. The resulting 128-bitsequence, considered as the coefficients of a polynomial of degree 127,when divided by G(x), will always produce a remainder equal tox⁷+x⁶+x+1. The input bits are unscrambled. Because all fields covered bythe HCS are transmitted at 2 MBaud and 2 bits per Baud, as describedbelow, these fields should be received correctly in many cases where thepayload is received in error. The HCS may be used in conjunction withsoft-decision error statistics to determine with high probabilitywhether the header was received correctly. This knowledge may be usefulfor optimizing the performance of ARQ and/or rate negotiation algorithmsdescribed below.

Returning to FIG. 6, it can be seen that the bit fields starting withthe destination address (DA) field 628 and ending with the FCS field 630are identical to the corresponding fields described in IEEE Std 802.3 asdepicted in FIG. 10 and are referred to as Link-level Ethernet Frame(packet) 632. The bits of a PHY-level Ethernet frame have an Ethernetpreamble 634 and start-frame-delimiter (SFD) 636 bits prepended to theLink-level frame, these bits are not present in the frames of thepresent embodiment. It is intended that IEEE assigned Ethernet MACaddresses are used for Destination Address (DA) 628 and Source Address(SA) 638. The Ethernet frame consists of an integer number of octets.Following Ether-type field 640 is Ethernet data field 642, FCS field 630and cyclic redundancy check (CRC) field 644. CRC field 644 is a 16-bitcyclic redundancy check computed as a function of the contents of the(unscrambled) Ethernet frame in transmission order, starting with thefirst bit of the DA field and ending with the last bit of the FCS field.The encoding is defined by the following generating polynomial.G(x)=x ¹⁶ +x ¹² +x ⁵+1Mathematically, the CRC value corresponding to a given frame is definedby the following procedure:

-   -   h) The first 16 bits of the frame in transmission order are        complemented.    -   i) The n bits of the frame in transmission order are then        considered to be the coefficients of a polynomial M(x) of degree        n−1. (The first bit of the Destination Address field corresponds        to the x^((n−1)) term and the last bit of the FCS field        corresponds to the x⁰ term.)    -   j) M(x) is multiplied by x¹⁶ and divided by G(x), producing a        remainder R(x) of degree <=15.    -   k) The coefficients of R(x) are considered to be a 16-bit        sequence.    -   l) The bit sequence is complemented and the result is the CRC.        The 16 bits of the CRC are placed in the CRC-16 field so that        x¹⁵ is the least significant bit of the first octet, and the x⁰        term is the most-significant bit of the last octet. (The bits of        the CRC are thus transmitted in the order x¹⁵, x¹⁴, . . . x¹,        x⁰.) The CRC-16, in conjunction with Ethernet's FCS, provides        for more protection from undetected errors than the FCS alone.        This is motivated by environmental factors that will often        result in a frame error rate (FER) several orders of magnitude        higher than that of Ethernet, making the FCS insufficient by        itself. For 4 MBaud payloads, a variable-length PAD field 646        follows CRC field 644 and consists of an integer number of        octets. The last octet of the pad field (PAD_LENGTH) specifies        the number of zero octets (0x00) preceding PAD_LENGTH. The value        of PAD_LENGTH must equal or exceed the number of zero octets        required to ensure that the minimum length of the transmission,        from the first symbol of the PREAMBLE64 through the last symbol        of the end of frame delimiter, is 92.5 microseconds. For 2 MBaud        payloads, there is no PAD field. The PAD field is not present in        a Compatibility Mode Frame, as described below. An example of a        compliant formula for generating PAD_LENGTH is max (102−N, 0),        where N is the number of octets from DA to FCS, inclusive. This        ensures that a collision fragment can be discriminated from a        valid frame by the transmission length detected by the carrier        sense function, as described below. The next field is End of        Frame (EOF) Delimiter field 648. The End-of-Frame sequence        consists of the first 4 symbols of the TRN sequence, or 0xfc        encoded as 2 bits-per-Baud at 2 Mbaud. This field is provided to        facilitate accurate end-of-carrier-sensing in low-SNR        conditions. A station demodulating a frame can use this field to        determine exactly where the last payload symbol occurred.

Turning now back to FIG. 5 and to FIG. 11, scrambler 520 is described inmore detail. Two difficult problems in CSMA/CD networks that useuncontrolled wiring (e.g. phoneline or powerline networks) areaddressed. The first problem is premature end-of-carrier detection andthe second problem is radio-frequency egress. With regard to prematureend-of-carrier detection, in powerline and phoneline CSMA/CD networks,there is a need to reliably detect the end of a frame in the presence ofsevere channel distortion. There also is a trade-off between the meantime required to detect the end of frame (from the actual end of frame)and the reliability of detection (probability of false alarm andprobability of missed detection). Two effects make end-of-framedetection difficult, particularly when the frame boundary detector isdecoupled from the demodulator:(1) the possibility of a long run ofinnermost constellation points, particularly for large constellations(e.g. high-order QAM or PAM); and (2) the possibility of a long run ofconstant or near-constant (nearest neighbors in a large constellation)symbols. A long run of innermost points can clearly be a problem if anenergy or matched filter detector is used to detect frame boundaries,and constant/near-constant symbol sequences (which produce tonaltransmitted sequences) are problematic because they may be highlyattenuated by the channel over which they travels Large constellationsare used in the system to achieve high spectral efficiency, and, hence,high data rates. Scrambling generally is an effective tool in combatingthese problems. However, there is still a non-zero probability of eithera sufficiently long run of innermost points or a sufficiently long runof constant/near-constant symbols to cause frame loss. Ordinarily, onewould not be concerned about these low-probability events. However,frame loss due to end-of-frame detection failure is deterministic:provided that the channel does not change between transmissions and theSNR is high, every transmission will fall victim to the same prematureend-of-carrier detection problem. With regard to radio frequency (RF)egress, in wired networks, there is always energy radiated from thewires that convey information. This egress can interfere with otherservices, some (e.g. amateur radio) which are specifically protected bygovernment agencies. In the “high-frequency” range, interference intoamateur radio receivers is a particular concern. These receiverstypically have a channel bandwidth of less than 3 kHz. While scramblingis also an effective tool for spreading packet energy over a wide band,thereby reducing the probability of harmful interference, many networkpackets (e.g. TCP acknowledgments) contain identical or nearly-identicaldata. Also, collisions between different stations' transmissions on thewire may result in the transmission of exactly the same data many timeswithin a short time window. Using the same scrambler seed (delay lineinitialization) for every transmitted frame may result in bursts ofnarrowband energy that are more likely to interfere with services suchas amateur radio. Therefore, in accordance with the present invention acommon solution to both problems is provided by two very simple circuitsfor mitigating this problem, one at the transmitter and another at thereceiver. Either an N-bit counter is implemented at the transmitter forevery active path (source+destination address combination) over which aframe may be sent on the network, or an N-bit pseudo-random numbergenerator is implemented. A simple linear-feedback shift register may beused to generate the pseudo-random number, if that approach is chosen.On every transmitted frame, the scrambler initialization circuit eithergenerates a pseudo-random N-bit number, or it increments the counter forthe path over which the frame will travel, modulo-2^(N). Eithertechnique is sufficient. The scrambler initialization circuit insertsthe N bits into any of the M (>=N) bits of a scrambler delay line. Notethat the N bits need not be contiguous in the M-bit delay line. For asufficient implementation, N>=2. The scrambler initialization circuitinserts same N-bit value into an unscramble part of the header of thetransmitted frame, so that the receiver may correctly recover thetransmitted bit sequence by initializing the descrambler with the chosenvalue. In one embodiment, the scrambler is the frame-synchronizedscrambler shown in FIG. 11, which uses the following generatingpolynomial: G(x)=X²³+x¹⁸+1. Bits 15 through 18 of a shift register areinitialized with a 4-bit pseudo-random number (or per-path countervalue). All other values are initialized to 1. The same value is placedin the unscrambled “SI” field of the Frame Control part of the header sothat the receiver may recover the chosen scrambler initialization. FIG.6 described above shows an example frame format which may convey the“SI” (scrambler initialization) bits to the receiver. FIG. 8 describedabove shows the components of the “Frame Control” field of the previousdiagram in this example. All bits up to and including “SI” in the areunscrambled in accordance with the present invention. Any bits followingthe SI field are scrambled using this technique.

Now to further describe the scrambler initialization aspects shown inFIG. 11, scrambler 520 is a frame-synchronized scrambler which uses thegenerating polynomial G(x)=X²³+x¹⁸+1. Bits 15 through 18 of shiftregister 650 are initialized with a 4-bit pseudo-random number. Thisvalue is placed in SI field 618 defined above in the order such thatregister position 15 is the MSB (bit 19 of frame control) and bit 18 isthe LSB (bit 16 of frame control). Scrambler 520 is bypassed during thepreamble bit field and the first 16 bits of Frame Control. Scrambler 520is initialized and enabled starting with the 17^(th) bit of FrameControl field 614. Scrambler 520 is bypassed after the last bit of theCRC-16 644, or the last bit of the PAD field 646, if present. The EOFsequence is not scrambled. The use of a pseudo-random initial scramblerstate results in a more uniform power-spectral density (PSD) measuredover multiple similar frames. This eliminates the problem of tones inthe PSD from highly correlated successive packets.

As can be seen in FIG. 11, input frame 5010 is the output of framing 510as seen in FIG. 5 which also generates SI value 618 as seen in FIG. 8.Bit values of 1 5002 are inserted into register bit locations 1–14 5004.Further bit values of 1 5006 are inserted into register bit locations19–23 5008. SI value 618 is inserted into bit locations 15, 16, 17, 185009 of register 650. Each of the additions are modulo 2, i.e., a bit,exclusive or, another bit, and so on. Input bits 5010 are exclusive or'dwith the output bits of register 650. Output bits 5012 are provided toconstellation encoder 530 as seen in FIG. 5., bit 1 being the mostrecent bit.

As to the descrambler initialization circuit, at the receiver, thedescrambler initialization circuit extracts the N bits of the “SI” fieldfrom the received frame. It then inserts the N bits into the samepositions of the descrambler delay line that were initialized in thescrambler, in the same order. (Note that the descrambler and scramblerdelay lines have exactly the same length, in bits.) In the exampleembodiment, all other bits in the descrambler delay line are set to “1”.The first bit inserted into the descrambler-is exactly the first bitinserted into the scrambler in the transmitter.

Turning again to FIG. 5, following scramber 520 is constellation encoder530. All bits up to and including the Ethertype field are encoded at 2MBaud, 2 bits per Baud. Starting with the 1^(st) bit following theEthertype field, the bits are encoded according to the PE field 624, upto the last bit of the CRC-16 644, or the last bit of PAD 646 if it ispresent. The EOF sequence 648 is encoded at 2 MBaud, 2 bits per Baud.Constellation encoder 530 performs bit to symbol mapping. The incomingbits are grouped into N-bit symbols, where N is the number of bits perbaud specified in PE field 624. The bit to symbol mapping is shown inFIGS. 12 a through 12 g. The symbol values are shown with bits orderedsuch that the right-most bit is the first bit received from scrambler520 and the left-most bit is the last bit received from scrambler 520.All constellations except for 3 bits-per-Baud lie on a uniform squaregrid, and all constellations are symmetric about the real and imaginaryaxes. The relative scaling of different constellations at a single baudrate is shown in FIG. 13. The constellation points are scaled such thatthe reference points have the values shown, with a minimum-distancetolerance of plus or minus 4%. The constellation points are scaled suchthat the outermost points have approximately equal magnitude. Symbols at4 MBaud are transmitted at 0.707 the amplitude of symbols at 2 MBaud. Ona transition from 2 MBaud to 4 MBaud, the first 4 MBaud symbol occur 0.5microseconds after the last 2 MBaud symbol. On a transition from 4 MBaudto 2 MBaud, the first 2 MBaud symbol occur 0.5 microseconds after thelast 4 MBaud symbol. This is illustrated in FIG. 14. If the number ofbits in a sequence at a given encoding rate (i.e. Baud rate and bits perBaud) is not an integer multiple of the number of bits per Baud, thenenough zero bits are inserted at the end of the bit-stream to completethe last symbol. The number of zero bits inserted is the minimum numbersuch that the length of the appended bit stream is an integer multipleof the number of bits per Baud. The number of octets in the originalinput bit stream can be determined unambiguously from the number ofsymbols transmitted. This is true because the maximum encoding size is 8bits per baud, which implies that the number of zero-bits appended mustalways be less than eight.

Referring again back to FIG. 5, complex symbols from constellationencoder 540 are input to QAM/FDQAM Modulator 540. QAM/FDQAM modulatorimplements Quadrature Amplitude Modulation (QAM). FIG. 15 shows anexample QAM implementation. The carrier frequency and transmit filtersare the same for Baud rates of 2 MHz and 4 MHz. Thus, a 2 MBaud signalis equivalent to an appropriately scaled 4 MBaud signal in which everyother symbol is zero. The QAM/FDQAM Modulator used in conjunction withthe present invention is described in more detail in the pendingapplication referenced above.

In addition to the frame formatting described above, the presentinvention provides for a Compatibility Frame format which is defined foruse by HPNA V2 nodes when they are sharing the phoneline with HPNA V1nodes. In this case, it is important that the V2 transmissions canmasquerade as valid V1 frames for correct carrier sense and collisiondetection behavior, even though the V1 nodes will not be able to recoverthe data from the frame. In this format, referring to FIG. 16, frame 700starts with a modified V1 AID field 710, followed by a V2 symbolsequence modified to have periodic gaps 720 so that a V1 receiver willdetect this signal as a series of pulses. The frame ends with trailer730 that includes 4-symbol V2 EOF 740 and a single pulse, EOP 750,generated by passing a QPSK symbol through the transmit path.

Referring to FIG. 17, transmitter aspect of PHY 320 operating inCompatibility mode is shown in functional black diagram form.Transmitter 800 includes framing 810 implementing the compatibility modeframing described above. Scrambler 820 is responsive to framing 810 andis identical to scrambler 520 described above in conjunction FIG. 5.Scrambler 820 is initialized at the same point in the frame controlfield. Coupled to scrambler 820 is Constellation Encoder 830Constellation encoder 830 is identical to the constellation encoder 530described above in conjunction with FIG. 5.

Referring back to FIG. 16, Preamble48 760 is defined as a repetition ofthree contiguous 16 symbol sequences (TRN16) that result from encoding0xfc483084 at 2 MBaud, 2 bits-per-baud, with scrambler 820 disabled. The72-symbol header, including frame control field (as defined in FIG. 7)and Ethernet DA, SA, and Ethertype fields, is contained within fourcontiguous subframes. It is scrambled and mapped to constellationpoints, prior to gapping, as previously described above. In the header,a subframe consists of: one non-information-bearing symbol (the leadsymbol), produced by Gap Insertion block 840, and 18 data symbols(header). A gap of 6 2-Mbaud zero symbols (silence) follows eachsubframe of 19 non-zero symbols. The format of each subframe and gap forthe 2-Mbaud header is depicted in FIG. 18. On subframes 0 and 2 of theheader, the lead symbol is defined as the first symbol of PREAMBLE48(bit sequence 00, encoded as QPSK). On subframes 1 and 3 of the header,the lead symbol is defined as the negation of the first symbol ofPREAMBLE48 (bit sequence 11, encoded as QPSK). The peak symbol amplitudeshown in FIG. 18 is defined hereinbefore in conjunction withconstellation scaling. The sign of the lead symbol alternates such thatthe output of the QAM/FDQAM modulator is the same at the beginning ofevery subframe. Negating the lead symbol of every other subframeaccounts for the 180-degree rotation introduced by the (7-MHz carrierfrequency) modulator and the odd number of symbols between the firstsymbols of two adjacent subframes. The special relationship between thecarrier phase of the first symbol of the preamble and of every leadsymbol is specific to the V2 compatibility mode. There are no suchrequirements in the V2 native mode.

Now turning to the 2-Mbaud and 4-Mbaud payloads in conjunction withcompatibility mode, the 2-Mbaud payload is encapsulated in subframes,consisting of: one non-information-bearing symbol (the lead symbol),produced by Gap Insertion block 840, between 1 and 18 data symbols(payload). A gap of 6 2-Mbaud zero symbols (silence) follows eachsubframe. On subframes 2*k, k>1, the lead symbol is defined as the firstsymbol of PREAMBLE48. On subframes 2*k+1, k>1, the lead symbol isdefined as the negation of the first symbol of PREAMBLE48. The firstfloor [N*8/(r*18)] subframes of the payload, where N is the number ofpayload bytes and r is the number of bits per baud, contain exactly 18information-bearing symbols. The last subframe of the payload containsthe remaining payload symbols, between 1 and 18. The last subframe isalso followed by a gap of 6 zero symbols. The format for all but thelast of the 2-Mbaud payload subframes and gaps is identical to theheader subframe and gap depicted in FIG. 18. For 3, 5, and 7 bits perbaud, the lead symbol is not a valid point in the constellation encoder.The 4-Mbaud payload is encapsulated in subframes, consisting of: onenon-information-bearing symbol (the lead symbol), produced by GapInsertion block 840, one zero symbol, and between 1 and 35 data symbols(payload). A gap of 13 4-Mbaud zero symbols (silence) follows eachsubframe. On subframes 2*k, k>1, the lead symbol is defined as the firstsymbol of PREAMBLE48. On subframes 2*k+1, k>1, the lead symbol isdefined as the negation of the first symbol of PREAMBLE48. The firstfloor [N*8/(r*35)] subframes of the payload, where N is the number ofpayload bytes and r is the number of bits per baud, contain exactly 35information-bearing symbols. The last subframe of the payload containsthe remaining payload symbols, between 1 and 35. The last subframe isalso followed by a gap of 13 4-Mbaud zero symbols. The format for allbut the last of the 4-Mbaud payload subframes and gaps is depicted inFIG. 19. The peak symbol amplitude and the amplitude of the othersymbols shown in the figure are defined above in conjunction with4-Mbaud constellation scaling. For 3, 5, and 7 bits per baud, the leadsymbol is not a valid point in the constellation encoder.

There are also two possible EOF/EOP sequences following a 2-Mbaudpayload and four possible EOF/EOP sequences following a 4-Mbaud payload.The EOF/EOP symbol sequence for the 2-Mbaud payload case is defined inthe table set forth in FIG. 20. P is the number of information-bearingsymbols in the last payload subframe and M is the number of payloadsubframes in the frame. The entire EOF/EOP sequence is encoded as QPSKat 2 Mbaud with the scrambler bypassed. The last symbol (similar to anV1 EOP) is used for accurate end-of-carrier timing in all V1 receivers.The EOF/EOP symbol sequence for the 4-Mbaud payload case is defined inthe table set forth in FIG. 21. P is the number of information-bearingsymbols in the last payload subframe and M is the number of payloadsubframes in the frame. The entire EOF/EOP sequence is encoded as QPSKat 2 M-baud with the scrambler bypassed. The last symbol (similar to anV1 EOP) is used for accurate end-of-carrier timing in all V1 receivers.

Referring back to FIG. 17, Modified AID Generator 850 is provided. Amodified V1 AID is prepended to every frame. The modified AID is definedas a V1 AID in which each pulse in the AID is replaced by a pulsedefined below. The AID number is one chosen by the sending station andconflicts are resolved by selecting a new AID number. The control wordalways indicates high-speed and low-power transmission. The use of theAID mechanism for collision detection implies that V2 has the samelimitation on the maximum number of nodes as V1 when in compatibilitymode. FIG. 22 shows the first part of a compatibility mode frame. Themodified AID pulse is generated by passing the first symbol of thePREAMBLE48 through the QAM/FDQAM modulator with the same initialmodulator phase as the first symbol of the PREAMBLE48. The modified AIDpulse is also used for the JAM sequence.

Referring back to FIG. 17 QAM/FDQAM Modulator operates continuously fromthe first symbol of PREAMBLE48, as described for QAM/FDQAM Modulator 540of FIG. 5.

Now turning to transmitter electrical characteristics, stations at aminimum are capable of transmitting and receiving 2 MBaud modulatedframes in native V2 frame format. In a preferred embodiment stations arecapable of transmitting and receiving 2 Mbaud Compatibility V2 frameformat. Stations at a minimum are capable of transmitting allconstellations from 2 bits-per-Baud to 8 bits-per Baud (PE values 1–7)and receiving all constellations from 2 bits per Baud to 6 bits per Baud(PE values 1–5). The R.M.S. differential transmit voltage does notexceed −15 dBVrms in any 2-msec window between 0 and 30 MHz, measuredacross a 135-Ohm load between tip and ring for any payload encoding. Thepeak differential transmit voltage does not exceed 580 mVpeak for anypayload encoding at either 2 Mbaud or 4 M baud. Stations that are nottransmitting emit less than −65 dBVrms measured across a 100-Ohm loadbetween tip and ring. The electrical characteristics described below asto spectral mask apply to both the V2 native mode and the V2compatibility mode. The V2 metallic power spectral density (PSD) isconstrained by the upper bound depicted in the FIGS. 23 a and 23 b withthe measurement made across a 100-Ohm load across tip and ring at thetransmitter wire interface. The mask applies to all payload encodings atboth 2 and 4 Mbaud. The resolution bandwidth used to make thismeasurement is 10 kHz for frequencies between 2.0 and 30.0 MHz and 3 kHzfor frequencies between 0.015 and 2.0 MHz. An averaging window of 213seconds used, and 1500-octet MTUs separated by an IFG duration ofsilence is assumed. A total of 50 kHz of possibly non-contiguous bandsmay exceed the limit line under 2.0 MHz, with no sub-band greater than20 dB above the limit line. A total of 100 kHz of possiblynon-contiguous bands may exceed the limit line between 13.0 and 30.0MHz, with no sub-band greater than 20 dB above the limit line. The 10 dBnotches at 4.0, 7.0 and 10.0 MHz are designed to reduce RFI egress inthe radio amateur bands. The mask is tested at PE values of 1 and 2 (2and 3 bits/symbol), as these payload encodings result in the maximumtransmitted power. The absolute power accuracy is +0/−2.5 dB relative to−7 dBm, integrated from 0 to 30 MHz. The passband ripple between 4.75and 6.25 MHz and between 8.0 and 9.25 MHz is less than 2.0 dB. Themagnitude of the V2 transmitter output is upper-bounded by the temporalmask shown in FIG. 24 for a compatibility mode pulse (the symbolresponse of the 2.0 transmitter). The response is measured across a100-Ohm load between tip and ring at the transmitter's WIRE interface.Output before t=0 and after t=5.0 microseconds is <0.032% of the peakamplitude. The first compatibility mode pulse in the modified AID isexactly the transmitter symbol response. The transmitter C-weightedoutput in the band extending from 200 Hz to 3000 Hz does not exceed 10dBrnC when terminated with a 600-Ohm resistive load. The transmitteremits no more than −55 dBVrms across a 50-ohm load between the centertap of a balun with CMRR >60 dB and the transceiver ground in the bandextending from 0.1 MHz to 50 MHz. The transmitter clock frequency isaccurate to within +/−100 ppm over all operating temperatures for thedevice. The minimum operating temperature range for this characteristicis 0 to 70 degrees C. In general, a +/−50 ppm crystal meets thischaracteristic. The R.M.S. jitter of the transmitter clock is less than70 psec, averaged over a sliding 10-microsecond window. The differentialnoise output does not exceed −65 dBVrms across a 100-Ohm load, measuredfrom 4 to 10 MHz with the transmitter idle. There is no gain or phaseimbalance in the transmitter, except with respect to constellationscaling as described above.

Now turning to a comparable receiver's electrical characteristics, thereceiver detects frames with peak voltage up to −6 dBV across tip andring at a frame error rate of no greater than 10⁻⁴ with additive whiteGaussian noise at a PSD of less than −140 dBm/Hz, measured at thereceiver. The receiver detects 1518-octet frames frames encoded as 2bits/symbol and 2 Mbaud with R.M.S. voltage as low as 2.5 mV at nogreater than 10⁻⁴ frame error rate. The R.M.S. voltage is computed onlyover time during which the transmitter is active. The receiver detectsno more than 1 in 10⁴ 1518-octet, 2 bits/symbol, 2 Msymbol/sec frameswith R.M.S voltage less than 1.0 mV. Both criteria assume additive whiteGaussian noise at a PSD of less than −140 dBm/Hz, measured at thereceiver, and assume a flat channel. The receiver demodulates frameswith payload encoded at 6 bits/symbol, 2 or 4 Mbaud (if implemented),and differential R.M.S voltage as low as 20 mV (measured over theheader) at a frame error rate less than 10−4 under the followingconditions: (1) White Gaussian noise with PSD less than −130 dBm/Hz isadded at the receiver, and (2) A single tone interferer with any of thefrequency band and input voltage combinations set forth in FIG. 25. Theapplied voltage is measured across tip and ring at the input to thetransceiver. The receiver demodulates frames with payload encoded at 6bits/symbol, 2 or 4 Mbaud (if implemented), and differential R.M.Svoltage as low as 20 mV (measured over the header) at a frame error rateless than 10−4 under the following conditions: (1) White Gaussian noisewith PSD less than −130 dBm/Hz is added at the receiver, differentialmode, and (2) A single-tone interferer, measured between the center tapof a test transformer and ground at the input to the transceiver, withany of the following frequency band and input voltage combinations setforth in FIG. 26. The common mode rejection of the test transformer usedto insert the signal should exceed 60 dB to 100 MHz.

The average return loss of the transceiver with respect to a 100-Ohmresistive load exceeds 12 dB between 4.75 and 9.25 MHz. Thischaracteristic applies to the transceiver powered on or in low-powermode (transmitter powered off). The average return loss with respect toa 100-Ohm resistive load exceeds 6 dB between 4.75 and 9.25 MHz with thetransceiver removed from a source of power. The magnitude of the inputimpedance is >10 Ohms from 0–30 MHz and conforms to the lower-bound maskset forth in FIG. 27. This characteristic applies to the transceiverpowered on, in low-power mode (transmitter powered off), or removed froma source of power. FIG. 28 shows an example of the input impedance of acompliant device with a lower bound mask.

With regard to the receiver aspects in accordance with the PHY layerprotocol, reference in made to FIG. 30, wherein receiver functionality900 is shown in block diagram form. Receiver functionality 900 performsthe reverse of that described above for transmitter 500, namely, uponreceiving a signal from 2–4 wire hybrid and performing front endprocessing as described in conjunction with FIG. 4 b, the followingoccurs: QAM/FDQAM Demodulator Gap Removal, Consellation Decoding,De-scrambling and De-framing, as is well-known in the art given theabove-defined transmitter functionality.

Homenetworking MAC Layer Overview

Now turning to the MAC Layer, the station media access control (MAC)function, as seen at the wire interface is described in more detail. TheHPNA V2 MAC is modeled after the carrier-sense multiple-access withcollision detection (CSMA/CD) MAC function of Ethernet (IEEE Std 802.3,1998 Edition), adapted to the V2 PHY and enhanced withquality-of-service (QoS) features. The MAC functions describedhereinbelow should not be confused with host interface and other layerfunctions typically implemented in a “MAC chip”. Also the MAC controlfunction should not be confused with IEEE 802.3 Clause 31 MAC Control.

Referring to FIG. 29, the MAC logical layers and functions are depicted.Although the MAC function is an essential part of the wire interfacecharacteristics, the system partitioning of PHY and MAC functions isimplementation dependent. In particular, it is intended that the presentembodiment can be implemented in an integrated PHY+MAC chip as well as aPHY-only chip that can be interfaced with a standard “MAC chip” usingthe Media Independent Interface (MII) described in IEEE Std 802.3-1998,clause 22.

When in Compatibility Mode V2 devices transmit either V1 Format framesor V2 Compatibility Format frames depending on the destination stationtype. The MAC operation in this mode is IEEE Std 802.3-1998 CSMA/CD MACwith BEB collision resolution and no access priority. When incompatibility mode the MAC operation is as specified in IEEE Std.802.3-1998, clause 4, for a MAC sublayer operating in half duplex modeat speeds of 100 Mb/s and below. The timing parameters to be used inCompatibility Mode are in accordance with the V1 PHY Specification,Version 1.1. In compatibility mode the MAC times the inter-frame gapfrom the de-assertion of the carrier sense signal, CAR_SENS. The timingof CAR_relative to the wire interface adheres to the timing specified inHPNA V1 PHY Specification rev 1.1, clause 3.3.

An implementation may have different individual CAR_SENS/MAC timingparameters provided the overall timing at the wire interface is the sameas CAR_SENS/MAC with the parameters specified. Further, In compatibilitymode the detection of collisions is as specified in HPNA V1 PHYSpecification rev 1.1, clause 2.5.3, with a JAM signal emitted asspecified in clause 2.5.4. ACCESS ID values are maintained as specifiedin clause 2.5.5.

Now turning to V2 Mode MAC Operation, each station on an V2 networksegment, when not in Compatibility Mode, executes the V2 MAC function tocoordinate access to the shared media. Switching between CompatibilityMode and V2 native mode is described hereinbelow. The MAC timingparameters for V2 Mode are also defined below.

The Carrier Sense Multiple Access/Collision Detect (CSMA/CD) mediaaccess method is the means by which two or more stations share a commontransmission channel. To transmit, a station waits (defers) for a quietperiod on the channel (that is, no other station is transmitting) andthen sends the intended message modulated as per the PHYcharacteristics. The transmission deferral is ordered by up to eightpriority levels, implementing absolute priority among stationscontending for access. If, after initiating a transmission, the messagecollides with that of another station, then each transmitting stationceases transmission and resolves the collision by choosing a BackoffLevel and defers to other stations that have chosen a lower BackoffLevel. The distributed algorithm for choosing Backoff Level guaranteesthat the access latency is tightly bounded. Each aspect of this accessmethod process is set forth in detail hereinbelow.

Referring again to FIG. 30, a transceiver functional block diagram of anembodiment of the present invention is shown which includes transmitfunctionality portion 500, counterpart receive functionality portion900, V1 compatability transmit and receive functionality portions 910,920, MAC functionality portion 1000 for both V1 and V2 modes, and 2–4wire hybrid portion 930. Included in MAC 1000 is carrier sensefunctionality portion 1100, collision detection functionality portion1200, and CSMA/CD collision resolution/rx frame synchronizationfunctionality portion 1300. Carrier Sense 1100 detects the starting andending times of a valid frame transmission on the wire. This is used todetermine when frames are present on the channel/transmission medium, aswell as being used to determine the presence of a Backoff Signal in aSignal Slot. Collision Detection 1200 detects the presence of a validframe transmission from some other station during an activetransmission, and for all stations, including non-transmitting stations,detects the received fragment that represents a transmission truncatedby a collision. Collision Resolution 1300 implements the distributedalgorithm that controls backoff. Although the performance of the blocksin the MAC function are implementation dependent, certain minimumperformance requirements are needed to ensure interoperability andcompatible sharing of the channel and are now described in more detail.

Referring to FIG. 31 a frame transmission that is valid with respect tothe specified Carrier Sense (CS) function (Valid CS Frame) is shown. Atransmitted Valid CS Frame will be affected by various signalimpairments when seen by any receiver. A Valid CS Frame at thetransmitter wire interface consists of: (1) A sequence of symbols whoseduration is equal to or greater than 92.5 microseconds (TX_FRAMEminimum) duration, but less than the maximum described below; (2) thefirst (64+16+24+24+8) symbols of which modulated at the Base Rate (2MBaud QPSK, 2 bits per symbol), where the initial 64 symbols consist ofthe preamble sequence 1110, where the next 64 symbol sequence (other)1120 is unique to the transmitting station, and where the next 8 symbolsare the (likely non-unique) bits of the Ethertype field; (3) anarbitrary Minimum Signal 1140, defined as a sequence of symbols whoseR.M.S. value over any 8-microsecond window shall never be more than 9 dBless than 100 mVrms across 100 Ohms (NOMINAL_RMS_VOLTAGE); (4) 4 symbolsof the EOF sequence 1150; (5) a trailing transient, whose peak voltagedoes not exceed 0.1% of the absolute peak transmitted voltage across a100-Ohm load at the WIRE interface at any point >5 microseconds afterthe last transmitted symbol of the EOF; and (6) a gap before the nexttransmission of this station of CS_IFG microseconds from the last symbolof the EOF to the first symbol of PREAMBLE of the next transmission,measured at the transmitter's wire interface. When a station detectswhat may be a collision it terminates transmission early, as describedbelow.

A Valid Collision Fragment at the transmitter wire interface consistsof: (1) a sequence of symbols of 70.0 microseconds (CD_FRAG) duration;(2) consisting of (64+16+24+24+8) symbols modulated at the Base Rate (2MBaud QPSK, 2 bits per symbol), where the initial 64 symbols consist ofthe preamble sequence, and where the next 64 symbol sequence is uniqueto the transmitting station, followed by 8 more symbols; (3) 4 symbolsof the EOF sequence; (4) a trailing transient, whose peak voltage doesnot exceed 0.1% of the absolute peak transmitted voltage across a100-Ohm load at the WIRE interface at any point >5 microseconds afterthe last transmitted symbol of the EOF; and (5) a gap of at leastCS_IFG+CD_FRAG microseconds from the first symbol of the PREAMBLE64 ofthe Valid Collision Fragment to the first symbol of the BACKOFF20 signalin the first Backoff Signal Slot (if present), measured at thetransmitter's wire interface. Receivers are only required to correctlydetect Valid CS Frames, Valid Collision Fragments, and the BackoffSignal described below. The Inter-frame Gap is 29.0 microseconds(CS_IFG), where the gap is defined at the points at which the previousframe drops below 50% of its peak and the current frame rises above 50%of its peak. Timing of subsequent transmissions following a Valid CSFrame or Valid Collision Fragment are based on a MAC timing reference,established by the receiver.

Referring to FIGS. 32 and 33, time following a transmission TX isdivided into slots: (1) an Interframe Gap (IFG)1400; (2) three BackoffSignal Slots 1500 (following collisions 1600); and (3) 8 priority slots1700. During these time periods the MAC is synchronized and the slottiming is defined by the rules for valid transmissions as set forthabove. After priority slot 0 there may be an arbitrarily long periodwith no transmissions followed by one or more stations attemptingtransmission. In this latter case the MAC is unsynchronized. When MACtiming is synchronized stations commence any transmission no earlierthan 0 and no later than 4 microseconds (TX_ON) after a slot origin,measured at the transmitter wire interface. Receiver Carrier Sensefunction 1100 as seen in FIG. 30 detects a maximum-amplitude Valid CSFrame over a range of 0 to at least 38 dB (CS_RANGE) flat-channelinsertion loss and additive noise with a flat PSD of −140 dBm/Hz at thereceiver with a missed frame rate of less than 10⁻⁴ and a prematureend-of-frame declaration rate less than 10⁻⁴. With additive whiteGaussian noise applied at the input with a PSD of −110 dBm/Hz, the falsecarrier detection rate is no greater than 1 per second. When the MAC isunsynchronized, the latest a station may commence transmission after apossible Valid CS Frame has appeared at the wire interface is 12microseconds (CS_DEFER) from the first symbol of the PREAMBLE64 of thedetected frame, as measured at the station's wire interface. CS_DEFER isthe maximum allowed carrier sense delay.

The V2 embodiment can be used for carrying media streams, such as videoand audio (as described in more detail below). To reduce the latencyvariation in these streams, a priority mechanism is implemented to allowhigher layers to label outgoing frames with priority, and guarantee thatthose frames will have preferential access to the channel over lowerpriority frames. The access priority method implemented is to delaytransmissions to a slot beyond the minimum inter-frame gap, based on thepriority level of the frame waiting to be transmitted. Referring back toFIG. 32, slots are numbered in decreasing priority, starting at priority7. Higher priority transmissions commence transmission in earlier slotsand acquire the channel without contending with the lower prioritytraffic. A station's Priority Slot is based on the PHY priority numberassociated with the frame ready for transmission (TX_PRI), as determinedby the network stack and communicated to the MAC. The station uses anyslot with a number less than or equal to TX_PRI, normally the slotnumbered exactly TX_PRI. FIG. 32 shows the relative timing of priorityslots. After priority slot 0 there are no more priority slots, and anystation with traffic at any priority level can contend on a first-come,first-served basis. All collisions after priority slot 0 are consideredto happen at PRI=0.) The Priority Slot width is 21.μ microseconds(PRI_SLOT). No station transmits in a Priority Slot numbered higher thanthe TX_PRI assigned to the frame being transmitted. Stations notimplementing priority default TX_PRI to a value of 1 when transmitting.Stations waiting for transmission monitor Carrier Sense, and defer if CSwas true prior to the start of the station's Priority Slot, or if beyondPriority Slot 0 the station defers if CS was true prior to the start oftransmission. Any station ready to transmit at the start of its PrioritySlot transmits if CS was false prior to the start of it's Priority Slot,without deferring if CS was asserted prior to the start of transmission.See FIGS. 34 a and 34 b, depicting transmission aspects, without andwith priority access, respectively. With priority access video trafficat priority level 7 gains access ahead of best effort traffic scheduledat level 1. The slot timer is restarted if there is some othertransmission that acquires the channel while a station is waiting at alower priority.

The TX_PRI value is the priority the MAC uses to schedule transmissionand is the value present in the PRI field of the frame header. Thisvalue is determined by a higher layer in the network stack. The PRIfield is used to transport the priority label from source todestination, to assist the destination in managing the receive queue.The 3 bit priority values referred to are “PHY priorities”. PRI=7 hasthe highest priority, PRI=0 has the lowest. There may be a mappingbetween PHY priorities and the Link Layer (LL) priority values asdelivered to the Link Layer by the NETWORK Layer. This mapping isdescribed herein below with regard to the Link Layer Protocols for V2.In general, the NETWORK layer or APPLICATION layer will determine whatpolicy is used to map traffic onto LL priorities. For instance, IETFIntegrated Services currently defines priority 0 as the default “besteffort” priority, and priority 1 as the penalty “worse than best effort”priority—and most implementations will map best effort to PHY PRI=1 andworse-than-best-effort to PHY PRI=0. The PHY priority mechanism isstrict priority (as opposed to schemes which allocate lower prioritiessome minimum percentage of network capacity)—higher priority trafficalways defers lower priority traffic. Higher priority traffic will belimited by admission control or other Link Layer policy mechanism toprevent over-subscription.

Two or more stations may begin transmitting in the same Priority Slotfollowing the IFG period. All stations monitor the channel to detect thecolliding transmissions of other stations. Colliding frame(s) will bereceived over a channel with impairments. FIG. 35, shows length ofcollisions and non-collisions. Passive stations can detect collisions byobserving the length of transmission fragment and the validity of thereceived PREAMBLE64. A Valid CS Frame is guaranteed to have a uniquesymbol sequence within the first 128 symbols (which are transmitted atBase Rate). The Ethernet MAC Source Address (SA) is used to guaranteeuniqueness. That field is scrambled, but the [scrambled SA, SI] tuplewill be unique. SI is the 4-bit scrambler initialization field, asdescribed above. After detecting a collision a station continues totransmit through the Ethertype field followed by an EOF sequence (symbol139) and then cease transmission. Thus, a station detecting a collisionwill cease transmission no later than 70.0 microseconds (CD_FRAG) afterthe beginning of the frame as measured at the wire interface. Theminimum size of a Valid CS Frame is 92.5 microseconds (TX_MIN). No jamsignal is transmitted on collisions. Passive stations, that are nottransmitting, monitor the length of Carrier Sense events and generate aCollision Fragment indication to the Collision Resolution function ifthe duration of carrier is less than 92 microseconds (CD_THRESHOLD).Stations do not recognize carrier events shorter than 32.0 microseconds(CD_MIN) as collisions. All transmitting and passive stations arecapable of detecting the collision of any maximum-amplitude Valid CSFrame transmission received over a range of 0 to 36 dB (CD_RANGE)flat-channel channel insertion loss and additive noise with a flat PSDof −140 dBm/Hz at the receiver with a missed-collision error rate ofless than 10−4 and a false collision error rate of less than 10⁻³, wherethe origin of the colliding frame is offset relative to the first symbolof the transmitted frame anywhere from earlier by up to 12 microseconds(CD_OFFSET_EARLY) to later by up to 15 microseconds (CD_OFFSET_LATE).Where there is a missed collision, the probability of detected andundetected errors in the payload data is enhanced, so CollisionDetection implementations are biased towards false collision errors,which are more innocuous.

A collision occurs when two or more stations are active with readyframes and are contending for access to the channel at approximately thesame time. Generally, collisions are between frames at the same prioritylevel. A distributed collision resolution (CR) algorithm is run whichresults in stations becoming ordered into Backoff Levels where only onestation is at Backoff Level 0 and can therefore acquire the channel.After the winning station completes its transmission, all stationsreduce their Backoff Level by one if it is greater than zero, and thenew station(s) at Backoff Level 0 attempt transmission. All stations,even those with no frame to transmit, monitor the activity on themedium. Also, the collision resolution cycle is closed, so that stationsthat did not collide are not allowed to contend for access to the mediumuntil all stations that collided have transmitted one frame successfullyor have forgone the right to transmit their waiting frame. Ultimatelyall stations that were contending for access in the initial collisiongain access to the wire and the collision resolution cycle is ended.This results in access latency being tightly bounded. This mechanismdiffers from Binary Exponential Backoff (BEB) used in other versions ofEthernet in that the Backoff Level does not determine the contentionslot chosen by a station—all stations at a given priority always contendin the slot corresponding to the access priority. Instead stations atnon-zero Backoff Levels defer contending until stations that are at zeroBackoff Level transmit. The method used is called Distributed FairPriority Queuing (DFPQ) as described in co-pending application Ser. No.09/0267,884, the content of which are expressly incorporated byreference herein. Each station maintains eight Backoff Level (BL)counters, one for each priority. The Backoff Level counters areinitialized to 0. The priority level of a collision can be inferred fromthe priority slot where the collision occurs. Consider the case wherestations are only contending on one priority. After a collision and anIFG, three special Backoff Signal slots (S0 . . . S2) are present beforethe normal sequence of priority contention slots occurs. Signal slotsonly occur after collisions, they do not follow successfultransmissions. Each active station pseudo-randomly chooses one of theslots, and transmits a Backoff Signal. More than one station cantransmit a Backoff Signal in the same slot. The active stations transmitBackoff Signals to indicate ordering information that determines the newBackoff Levels to be used. All stations (even those without a frameready to transmit) monitor collision events and the Backoff Signal slotsto compute the Backoff Level. If an active station sees a Backoff Signalin a slot prior to the one it chose, it increases its Backoff Level.Those stations at Backoff Level 0 (ones that are actively contending)that saw no Backoff Signals prior to the one they chose, remain atBackoff Level 0 and contend for transmission in the priority slot equalto TX_PRI that immediately follows the Backoff Signal sequence.Eventually, only one station remains at Backoff Level 0 and successfullygains access to the channel. Stations with higher priority waitingframes may pre-empt the collision resolution by transmitting in ahigher-priority slot. All stations, even those not contending for accessto the wire, also maintain a Maximum Backoff Level (MBL) counter perpriority, which is incremented for each Backoff Signal seen anddecremented when a successful transmission occurs. The MBL is non-zerowhenever a collision resolution cycle is in progress. When a stationfirst becomes active, if MBL is non-zero, BL is initialized to contents[MBL], otherwise BL is initialized to 0. This ensures that all currentlyactive stations gain access to the channel before stations can re-enterthe waiting queue. The BACKOFF20 signal is a symbol sequence consistingof 16 symbols of the preamble sequence (TRN16) transmitted, followed bythe 4 symbol EOF sequence. Detection of the BACKOFF20 signal(s) in aBackoff Signal slot must be possible even if more than one stationselects the same slot. Stations implement saturating 4-bit BL and MBLcounters. The width of the Signal Slot is 32 microseconds (SIG_SLOT).Stations implement the MAC function with collision resolution whosebehavior matches the procedural model described below.

The procedural model uses a pseudo-code modeled after Concurrent Pascal.IEEE Std 802.3 1998 Clause 4.2.2 provides an overview of thispseudo-code. The code set forth at the end of the specification modelsthree independent concurrent processes (Deference, Transmitter,Receiver), which interact through shared variables. The Deferenceprocess is driven by the detection of transmissions on the channel, andtimes the boundaries for Signal Slots and Priority Slots. The sharedvariable current Priority signals the Transmitter process when atransmission slot exists.

Referring to FIG. 36, certain Mac parameters are set forth. Where atolerance is indicated, Δ=63 nanoseconds. The Link-level frame consistsof the DA through FCS fields, prior to the PHY-level frameencapsulation. All V2 stations transmit link-level frames with a minimumof 64 octets. The payload field of link-level frames smaller thanminFrameSize is padded with any value octets appended after the suppliedpayload to make the frame minFrameSize long. The maximum standardEthernet frame is 1518 octets, but some V2 link-layer encapsulations mayadd additional octets. All V2 stations are able to transmit and receivelink-level frames with up to 1526 octets. No V2 station transmits linklevel-frames with more than (PE+1)*1024 octets for 2 MBaud and not morethan (PE−7)*2048 octets for 4 MBaud. The number of octets specifiedcounts DA through FCS, and does not count preamble, header, CRC-16 orPAD or EOF. This will result in a maximum frame duration of 4166microseconds for a frame with PE=15. A V2 station defaults the maximumlength frame it will send to a given DA to 1526 octets until it candetermine that the receiver can support larger transmission units (e.g.by use of the CSA announcement of CSA_MTU as described below with regardto Link protocols. These maximums establish an upper bound on theduration of a given transmission and an upper bound on the maximum framesize that receivers must accommodate.

Homenetworking Link Layer Overview

In accordance with the present invention the following link controlfunctions are implemented: (1) Rate Negotiation; (2) Link Integrity; (3)Capability Announcement; (4) Limited Automatic Repeat reQuest (LARQ).These link functions use control frames to carry protocol messagesbetween stations. V2 includes a standardized mechanism for Link Layernetwork control and encapsulation. Control frames are data link layerframes that are identified by IEEE assigned Ethertype value (0x886cdesignated for the Assignee of the present application) in theType/Length field of the frame, and further distinguished by individualsub-types. The link control entities may be implemented in hardware ordriver software. Link Control frames are not seen by layer 3 (IP) of thenetwork stack, and are not bridged between network segments.

It should be noted that the Minimal Link Protocol Support Profile forHPNA V2 Link Protocols embodiments of the present invention allow lesscomplex implementations of the HPNA V2 characteristics. While each ofthe four control protocols serves an important function in the operationof the network, it is possible to implement minimal support for CSA andLARQ that is compatible with fully functional implementations and doesnot detract from the overall performance of other stations. The shortername, Minimal Profile, will be used hereinbelow. Full support of all thelink protocols, called the Full Link Protocol Support Profile or FullProfile for short, is assumed unless Minimal Profile is explicitlymentioned.

Referring to FIGS. 37 and 38, there are two basic formats for a LinkControl Frame, a long subtype and a short subtype. The long subtypeformat is provided for future specified control frames where the amountof control information exceeds 256 octets. The control and encapsulationframes described herein use the short subtype format. In the frameformats defined below, note that before transmission the Link ControlFrame is converted into a physical layer frame by adding Preamble, FrameControl, CRC-16, PAD and EOF as described above.

Describing first the short format as shown in FIG. 37, the SSVersionfield is recommended for all protocols using the Short Form Link ControlFrame header, and specifies which format version of the controlinformation is used. This allows future extension of each SSType.SSLength is checked only to ensure that enough control information ispresent. New, backwards compatible, frame formats may contain additionalfixed data fields, but will always contain the fixed fields specified inearlier formats, so protocol implementations simply use the latestversion that is less than or equal to SSVersion. The Next Ethertypefield is implemented for all SS headers. Among other things, it supportsbackward compatibility by enabling receivers to always strip shortformat link layer headers. If the Next Ethertype field is zero, then theframe is a basic control frame and is dropped after processing thecontrol information it contains. The Next Ethertype is always the lasttwo octets of the control header. The position of Next Ethertype in theframe is determined using the SSLength field in order to ensure forwardcompatibility. V2 receivers are able to remove at least oneencapsulating header with an unknown subtype from any received dataframe. Future embodiment versions may require the processing of multipleheaders, such as might occur if a Rate Request Control Frame wereinserted into (piggybacked on) a regular data frame with a LARQ header.The header and trailer for standard Ethernet frames are cross-hatched inFIG. 37, in order to highlight the formats of the control informationframes.

Describing now the long format as shown in FIG. 38, an LSVersion,similar to SSVersion, is recommended for all Long Format subtypes. ANext_Ethertype field is implemented for all Long Format subtypes. Ingeneral, if Long Format subtypes are not understood by the receiver (afact possibly announced via future CSA options) then they are dropped.Processing requirements with respect to forward compatibility, droppingof unknown frame types with Next_Ethertype=0, and removal of Long Formatheaders with Next_Ethertype !=0, are identical to those for Short FormatControl Frame headers. Network transmission order of frame fields isfrom the top to the bottom of each figure. Within a field, the MSByte ofthe field is the first octet of the field to be transmitted, with theLSBit of each octet transmitted first. Subsequent bytes within a fieldare transmitted in decreasing order of significance. When subfields arepictured in the figures, the ordering shown is decreasing significancefrom the top to the bottom of the figure.

The payload encoding (PE) that can be achieved is a function of thechannel quality between source and destination, and the channel qualitygenerally differs between each pair of stations depending on the wiringtopology and specific channel impairments. Therefore the RateNegotiation function in a destination station uses Rate Request ControlFrames (RRCF), as shown and defined in FIG. 39, to provide informationto a source station as to the payload encoding that the source stationuse to encode future frames sent to this destination, and to generatetest frames to assist a receiver in selecting the most appropriate bandto use. The policy which the destination station uses to select thedesired payload encoding and the policy it uses to decide when totransmit Rate Request Control Frames are implementation dependent.Algorithms for selection of payload encoding and time to transmit RRCFfor a preferred embodiment are set forth in more detail below. Stationsavoid transmission policies that can result in excessive RRCF traffic.The PHY payload modulation can use 2 to 8 bits-per-Baud constellationsand one of several defined bands which are combinations of Baud rates,modulation type (QAM vs. FDQAM) and carrier frequency, as describedabove. The RRCF specifies a maximum constellation (bits per Baud) thatthe receiver (ReqDA) wishes to be used in a given band, or indicatesthat a given band is not supported. Additional bands may exist in futureversions, and can be described with band descriptors {PE, rank} addedafter Band 2. If additional bands are present, their descriptors willappear between Band2_Rank and RefAddr1, and HPNA V2 stations take theirpresence into account when determining the location of the RefAddr list.In FIG. 39, the cross-hatched fields will not always be present. V2stations ignore band characteristics beyond Numbands=2. If a receiverdoes not specify a band in an RRCF, or specifies a PE of 0 for a band,then transmitters do not use that band. The NumBands and NumAddr fieldsare placed next to each other so that all the fixed fields can bereferenced at known offsets in the frame. FIG. 40 shows the assignedvalues that may appear in the band description entries in the RateRequest Control Frame. FIG. 41 shows the values that may appear in theOpCode entry in the Rate Request Control Frame. FIG. 42 describesfurther terms and definitions. Rate Negotiation is defined over simplexlogical channels. A separate channel is defined for each combination ofEthernet DA and SA. There is no explicit channel setup procedure. A newchannel is implicitly defined when a packet is received from a new SA orsent to a new DA. Each channel has a single sender but can have multiplereceivers. Receivers operate independently. Rate control frames (allOpCodes) are sent with a priority corresponding to Link Layer priority7.RRCF are not sent with a Link Layer priority of 6. RRCFs may be sentwith a lower Link Layer priority, from the set [5,4,3,0]. However, theLink Layer priority of an RRCF is not lower than the highest Link Layerpriority received in the last 2 seconds from the station to which theRRCF is being sent. Rate Change Requests (OpCode=0) is sent with anencoding of 2 Mbaud FDQAM at 2 bits per baud (PE=1) when the network isoperating in V2 mode, and with V1 encoding when the network is operatingin Compatibility mode. Selection of the encoding for Rate Test Requestframes and Rate Test Reply frames is described below. Each stationmaintains a timer with a period of 128 seconds. There is no attempt tosynchronize this timer between stations. The timer is not modified byreceipt or transmission of any frames. The timer interval is used whendetermining which nodes have been actively sending to multicast andbroadcast addresses and when sending reminder RRCFs in reference tomulticast and broadcast addresses.

Turning now to the sender operation, the logical channel stateinformation is accessed to determine the sender PE to use fortransmission. The channel is created if necessary, and the sender PEdefaults to PE=1 (2 bits-per-baud, 2 Mbaud FDQAM) if the network is inV2 mode, or to PE=8 if the network is operating in Compatibility mode.Logical channel state information includes the node type (if known), thesender PE and the receiver PE for each band for which this informationhas been specified. When first entering Compatibility mode, alltransmissions to all nodes are sent with V1 encoding for a period of 60seconds, regardless of the sender PE associated with the logical channelstate information. While in Compatibility mode, if the logical channelstate information for any active V2 node includes a value of senderPE=8, then all transmissions to all nodes are sent with V1 encoding.When no active V2 node's logical channel state information includes avalue of sender PE=8, transmissions revert to the rate specified by thesender PE associated with each channel. While in Compatibility mode, allmulticast and broadcast frames are always transmitted with V1 encoding.While in V1 mode, all frames are always transmitted with V1 encoding.For each of the RefAddrs in the RRCF (starting with RefAddr0, the SA ofthe RRCF frame), the logical channel state information is accessed, ifany exists, corresponding to the RefAddr, and the sender PE is updatedaccording to the band characteristics in the RRCF. If no logical channelstate information exists for RefAddr0, the station creates a new logicalchannel state entry and initializes the sender PE according to the bandcharacteristics in the RRCF. If no logical channel state informationexists for additional RefAddrs, the station may either ignore thoseaddresses or create new logical channel state entries and initialize thesender PE according to the band characteristics in the RRCF. Formulticast addresses and the broadcast address, senders use a rate thatis receivable by all nodes actively listening to that address. Senderstations may enforce a minimum PE which they will use to transmit to agiven multicast channel, based on application-level information aboutQuality of Service (QoS). It is desirable to send at the highest ratesupported by the channel. Hence, if a RefAddr is a multicast address orthe broadcast address, the sender uses the PE value which yields thehighest raw bit-rate, but which is not greater than any of the bandcharacteristicss provided by the nodes actively listening to thataddress. Active multicast listeners are defined as any stations whichhave, in either of the last two 128-second intervals, either (1) sentany frame to the multicast address or (2) sent a RRCF to this stationwith the multicast address listed in the RefAddr list. Active broadcastlisteners are defined as any stations which have, in either of the lasttwo 128-second intervals, either (1) sent any frame (with the optionalexception of Link Integrity frames using PE=1) to the broadcast addressor (2) sent a RRCF to this station with the broadcast address listed inthe RefAddr list. For each supported band encoding a Rate Test Replyframe (RRCF OpCode 2) is generated to the requester encoded using thespecified payload encoding. The contents of the RRCF is the currentlogical channel state info. Support for Rate Test Request frames is onlyrequired in stations that implement additional bands beyond Band1.Stations that only implement Band1 silently discard received Rate TestRequest frames. An active V2 node is any station from which a frame hasbeen received in either of the last two 128-second intervals. Whenever atransition to Compatibility mode occurs, sender PE is reset to a valueof 8 for all channels. Whenever a transition to V2 mode occurs, senderPE is reset to a value of 1 for all channels. A station which is notcapable of transmiting or decoding compatibility mode frames (hereafterreferred to as a non-compat station), adheres to the followingadditional rules. Whenever a non-compat station transitions from V2 toCompatibility mode, it transmits a broadcast RRCF (i.e. an RRCF withDA=“FFFFFFFFFFFF”) requesting PE=8 for all applicable refAddr values,within the first 30 seconds after the mode transition. As long as thenon-compat station remains in Compatibility mode, it continues totransmit broadcast RRCFs requesting PE=8 for all applicable refAddrvalues, at a rate of one RRCF every 128 seconds. These RRCF transmissionrules replace all other RRCF transmission rules for a non-compat stationduring Compatibility mode.

Now turning to receiver operation, the following baseline algorithm forlimiting the number of RRCFs is a preferred embodiment. Alternativeimplementations do not generate more RRCFs than the suggestedimplementation. Nodes that are interested in participating in amulticast or broadcast channel provide a mechanism to ensure that allsources of frames sent to the multicast address of the channel arereminded of this node's participation in that channel at least onceevery 128 seconds. For each channel, a Rate Control Backoff Limit (RCBL)is maintained that ranges in value from 1 to 1024, and a Rate ControlBackoff Frame Count (RCBFC). RCBL is initialized to 1, and RCBFC isinitialized to 0. For each received frame, the new desired PE iscomputed. A sample algorithm for selection of desired PE is describedbelow. If the new desired PE is different from the previous value of thedesired PE, then RCBL is reset to 1, and RCBFC is reset to 0. The newvalue is saved for desired PE. If the PE of the received frame isdifferent from the new desired PE, then RCBFC is incremented by 1. IfRCBFC is now greater than or equal to RCBL, then an RRCF is sent to thesource of the frame, RCBFC is reset to 0, and RCBL is doubled up to amaximum of 1024. If a multicast channel is active (based on receivingframes other than RRCFs within the last two 128-second intervals), andmore than 128 seconds have passed since the receiver has sent a frame tothis multicast address, an RRCF is transmitted with the current receiverPE to any nodes that have sent frames to that multicast address, with aRefAddr set to the multicast address in question. Multiple multicastaddresses may be aggregated into a single RRCF being sent to a node thathas been active on multiple multicast addresses. However, only addressesfor which the intended recipient of the RRCF has been active areincluded. In RRCF messages, requesting stations attempt to specify themaximum payload encoding that they believe will have an acceptable errorrate, in order to maximize the aggregate throughput of the network. At aminimum, the 2 MBaud band is specified in an RRCF. An example algorithmsuitable for use by devices implementing a single band (Band1) onnetworks with additive white noise and impulse noise is now described.Other algorithms are possible which may better optimize the selectedpayload encoding based on the measured channel conditions. For eachimplementation, a table of average slicer mean squared error (ASMSE)required for each payload encoding (except PE=8) to achieve a packeterror rate (PER) of 1e−3 is compiled. This table is defined asDOWN_LARQ. A second table is defined with a target PER of 1e–6. Thistable is defined as DOWN_NOLARQ. UP_LARQ is defined as DOWN_LARQ withall ASMSE values decreased by 2 dB and UP_NOLARQ is defined asDOWN_NOLARQ with all ASMSE values decreased by 2 dB.

The following steps describe how to select the new payload encodingdesired for a particular channel, (new_pe), given the current payloadencoding desired on that channel, (curr_pe), and a new frame is receivedon that channel.

-   1. Keep a history—window of 16 HPNA V2 frames per channel. For each    channel, compute the ASMSE over all frames in the history window    that did not have a CRC error.-   2. If in Compatibility mode, assess whether or not enough margin    exists in the system to allow proper detection of compatibility    frames on a per channel basis. If, for any given channel, such    margin is determined not to exist, then set new_pe=8 for that    channel. If such margin is determined to exist and curr_pe=8, set    new_pe=1. If such margin is determined to exist and curr_pe 8, set    new_pe=curr_pe. If new_pe=8 or curr_(—l pe=)8, then exit. Else:-   3. If all the frames in the history window were received with a CRC    error, set new_pe=1 and exit. Else:-   4. If LARQ is in use on a channel, find the greatest payload    encoding in the UP_LARQ table with an ASMSE greater than or equal to    the ASMSE computed in step 1. If LARQ is not in use, use the    UP_NOLARQ table. Define this payload encoding as new_up_pe-   5. If LARQ is in use on a channel, find the greatest payload    encoding in the DOWN_LARQ table with an ASMSE greater than or equal    to the ASMSE computed in step 1. If LARQ is not in use, use the    DOWN_NOLARQ table. Define this payload encoding as new_down_pe-   6. If new_up_pe>curr_pe, set new_pe=new_up_pe and exit. Else:-   7. If new_down_pe<curr_pe, set new_pe=new_down_pe and exit. Else:-   8. If neither 6 nor 7 is satisfied, set new_pe=curr_pe.    The offset between the up and down rate selection tables provides    the algorithm with hysteresis to provide stability in selection of a    payload encoding in the presence of minor variations in ASME. Due to    this offset, conditions 6 and 7 cannot both be satisfied    simultaneously. The combination of the 16 frame history window with    the selection hysteresis prevents the rate selection algorithm from    generating an excessive number of rate changes while remaining    responsive to significant changes in the channel conditions. The    selection algorithm for the value PE=8 in step 2 should also include    hysteresis to avoid generating an excessive number of rate changes    while remaining responsive to significant changes in the channel    conditions. Periodically, but at a rate not to exceed once every 128    seconds (except as described below) a receiver may send a Rate Test    Request frame to a sender to test if the channel can support a    different band. The band encodings represent the encodings for which    the receiver would like the sender to generate test frames. NumAddr    is set to 0 in Rate Test Request frames. Rate Test Request frames    are sent encoded at the current negotiated rate for the channel from    the receiver to the sender. Support for Rate Test Request frames is    only required in stations that implement additional bands beyond    Band1. Stations that only implement Band1 need not provide a    mechanism for generating Rate Test Request frames. Upon receipt of a    Rate Test Reply frame, the receiver uses the demodulation statistics    for this frame, and any previously received Rate Test Reply frames    using this encoding, to make a decision as to the channel's    capability to support the tested band encoding. If the decision is    that the channel is not capable of supporting the tested band    encoding, the receiver does not generate another Rate Test Request    frame for at least 128 seconds. If the decision is that the channel    is capable of supporting the tested band encoding, the receiver may    repeat the test to collect more data, at a maximum rate of one    RateTest Request frame every second, with a maximum of 16 additional    tests. At this point, the receiver generates a Rate Change Request    to the sender specifying the new band encoding. Support for Rate    Test Reply frames is only required in stations that implement    additional bands beyond Band1. Stations that only implement Band1    silently discard received Rate Test Reply frames. Whenever a    transition to Compatibility mode occurs, the receiver PE is set to a    value of 8 for all channels for a period of 60 seconds, to match the    Sender nodes' behavior. Whenever a transition to V2 mode occurs, the    receiver PE is set to a value of 1 for all channels.

The Link Integrity Function is now more fully described. In addition tothe implementation aspects set forth hereinbelow, the concepts set forthin related U.S. patent application Ser. No. 09/619,553 entitled “AMethod And Apparatus For Verifying Connectivity Among Nodes In ACommunication Network”, which is incorporated herein by reference. Thepurpose of the Link Integrity Function is to provide a means forhardware and/or software to determine whether or not this station isable to receive frames from at least one other station on the network.In the absence of other traffic, a station periodically transmits a LinkIntegrity Control Frame (LICF) to the Broadcast MAC address, with theinterval between such transmissions governed by the method describedbelow.

When in Compatibility Mode, a V2 station transmits LICF's once persecond as HPNA V1 frames. The standard frame format defined below isused (including the use of Broadcast (0xFFFFFFFFFFFF) for thedestination MAC address). When in native V2 mode, all stations implementthe following function to ensure that, with high probability, within any1 second interval there is either (1) at least one LICF sent to theBroadcast MAC address from this station, or (2) at least one packetaddressed to the Broadcast MAC address received from each of at leasttwo other stations. Additionally, all stations send at least one LICFevery 64 seconds. In V2 mode, a Link Packet is any frame received with avalid header FCS. In compatibility mode, a Link Packet is any V2compatibility frame with valid header FCS or any V1 frame fragment witha valid AID header and a PCOM field. Each station maintains afree-running timer with a period of 1 second. There is no attempt tosynchronize this timer between stations. The timer is not modified byany link state transitions or by the reception of any frames. This timeris the source of the timeout event used in the link integrity statetable described below. Each station maintains a 6 bit FORCE_SEND counterwhich is initialized to a random value between 30 and 63. Thisinitialization value may be selected once at node startup and used foreach re-initialization of the FORCE_SEND counter, or a new random valuemay be selected for each re-initialization of the FORCE_SEND counter.Each station has a register (SA1) that can be set from the SA of areceived Link Packet. When in native V2 node, an LICF is sent with apriority corresponding to Link Layer priority 7. The PE for an LICF isdetermined by accessing the RRCF logical channel information for thebroadcast channel. Each station sends a Link Integrity Control Frame(LICF) according to the state diagrams shown in FIGS. 43 a and 43 b. Thestate diagram depicted in FIG. 43 a gives a pictorial view of the statetransitions, with some minor loss of detail, including omission ofevents that do not cause state transitions (and have no associatedactions), and the collapsing of multiple events into a single transitionwith a more complex description of the action. FIG. 43 b provides acomplete state table, with associated actions. The timeout event is theperiodic expiration of a onesecond free-running timer. Initial State:DOWN, Force_Send initialized: 30<=Force_Send<=63. The FSM is a unifiedimplementation which provides the required behavior in both native V2and Compatibility compatibility modes. Link Integrity Status isindicated when in any state but DOWN. In a preferred embodiment allstations include a visible Link Status Indicator (LSI) (e.g. an LED) forindicating Link Integrity Status. FIG. 44 shows a Link Integrity ShortFrame.

The Capability and Status Announcement aspect in accordance with thepresent invention is now described more fully. A mechanism is definedfor network-wide negotiation, capability discovery and statusannouncement. It is based on periodic broadcast announcements, calledCapabilities and Status Announcements (CSA) sent in CSA Control Frames(CSACFs). The defined status flags allow determination of the station'sHPNA version, optional feature support, and link-layer priority usage,as well as communication of network configuration commands. The purposeof the protocol is to distribute to all stations the complete set ofstatus flags in use on the network, so that stations can makeoperational decisions based on those flags with no further interaction.Stations use the CSA Control Frame as shown in FIG. 45 and the CSA Flagdefinitions as shown in FIG. 46. Stations send a CSA Control Frame onceper minute or when a change in the station's current status requires theannouncement of new (or deleted) flags. A station sending a CSA ControlFrame announcing a status change sends a second copy of the most recentCSACF a short interval after the first, since it is always possible tolose a frame due to temporary changes in the channel, impulse noise,etc. The interval is randomly selected (not simply fixed), and chosenfrom the range 1 to 1000 milliseconds, inclusive. CSA Control Frames aresent with a priority corresponding to Link Layer priority 7. CSA ControlFrames are always sent to the Broadcast address (0xFFFFFFFFFFFF). The PEfor a CSA control frame is determined by accessing the RRCF logicalchannel information for the broadcast channel. A Request op-code isdefined to allow a station to quickly gather complete information aboutall stations. Upon receiving a CSA control frame with the Requestopcode, a station transmits a current CSA message after a delay of ashort interval, using the same mechanism (and parameters) that delaysthe second copy of CSA announcements, described above. Referring to FIG.45, the first three fields beyond the Ethernet header comprise thestandard header for short format control frames. Referring to FIG. 46,flags are used for CSA_CurrentTxSet, CSA_OldestTxSet, andCSA_CurrentRXSet in Capabilities and Status Announcement control frames.Thirty-two bit-flags are supported for announcing status andconfiguration information. The flags are divided into three basicgroups: mode selection flags including HPNA version information,supported options, and in-use TX link layer priority announcements.These flags are added to the global state as soon as announced, andremoved when no longer announced by any station, either through explicitdeletion or by timing them out. An in-use TX link layer priority will beannounced for a period of one to two minutes after the last frameactually sent with the priority, until the aging mechanism causes it tobe deleted from CurrentTxSet. The default set of status flags, used toinitialize the NewTxSet (defined below), is defined to be the priorities0 and 7, the station's HPNA version, and any supported options. Thebasic time interval used to age out non-persistent status information isone minute. Each station has a repeating timer set to this interval. Thetimers in different stations are not synchronized, and synchronizationis avoided. The description below refers to the time between oneexpiration of this timer and the next as a “period”. The “current”period refers to the time since the most recent expiration of the timer.A CSA frame is sent at the end of each interval. FIGS. 47 and 48 depictvariables and timers respectively. Each station maintains five basicsets of status and priority information, as shown in FIG. 49. Inaddition, three more composite sets are defined as the union of two ormore of the basic sets, as shown in FIG. 50. The composite sets are keptin sync with their component basic sets.

Now turning to the Capabilities and Status Announcement Protocoloperation, the CSA Protocol does not directly process transmit frames.When the LARQ protocol as discussed below is in use (Full Profilestations), CSA looks at the LL priority of the frame as it wouldnormally be sent to the driver: 1. If the LL priority is not already inNewTxSet, it is added to NewTxSet. 2. If the LL priority was not alreadyin NewTxSet and it is not in PreviousTxSet, then a new CSA control framewith the CSA_Opcode set to 0 (Announce) is sent , and theRetransmitTimer is started. If the timer was already running, it is thencanceled and restarted. The current PHY priority mapping function forthe driver is updated. The receiver may want to save a copy of some orall of the most recent CSA from each other station as a simple way oftracking other station's capabilities and status:

-   1. The status and options flags from the CSA_CurrentTxSet are    recorded (optionally) in a table indexed by the SourceMacAddress.    The options flags are used to select use of optional functions    between pairs of stations that implement the same options.-   2. If the CSA_Opcode in the frame is 1 (Request), then the    RetransmitTimer is started. If the timer is already running it is    recommended that it be left running, although this is not required    and cancellation followed by restart is allowed.-   3. If CSA_CurrentTxSet has a flag not already in NewRxSet, then the    flag is added to NewRxSet, and it is determined whether this flag is    not present in the PreviousRxSet. The corresponding boolean    expressions are as follows:    -   NewRxFlags=(CSA_CurrentTxSet & ˜NewRxSet)    -   NewRxSet|=NewRxFlags    -   ReallyNewFlags=NewRxFlags & ˜(PreviousRxSet|CurrentRxSet)-   4. CSA_OldestTxSet is compared with CSA_CurrentTxSet. If a flag has    been deleted, and if that flag is also missing from    CSA_CurrentRxSet, the flag is then deleted from NewRxSet, and    PreviousRxSet. The corresponding boolean expressions are as follows:    -   DeleteSet=(CSA_OldestTxSet & ˜CSA_CurrentTxSet) &        ˜CSA_CurrentRxSet    -   NewRxSet=NewRxSet & ˜DeleteSet    -   PreviousRxSet=PreviousRxSet & ˜DeleteSet-   5. If either ReallyNewFlags or DeleteSet are non-zero, then the    network mode and priority mapping are updated, as necessary. When a    CSP_Timer timeout occurs, the various status sets are rolled over,    the composite sets are re-computed, and a CSA is sent. The    RetransmitTimer, if needed, is set.    -   1. Move NewRxSet to PreviousRxSet.    -   2. Set NewRxSet to 0 (empty set).    -   3. Move PreviousTxSet to OldestTxSet.    -   4. Move NewTxSet to PreviousTxSet.    -   5. Set NewTxSet to the default set, consisting of this station's        highest supported version, current configuration flags if any        (normally none), currently supported options, and the default        priority set {0,7}.    -   6. Update CurrentTxSet, CurrentRxSet, and InuseSet (at least        logically, an implementation need not keep separate copies of        these values).    -   7. Send a CSA frame with the CSA_Opcode set to 0 (Announce),        including the updated flags.    -   8. If CSA_CurrentTxSet and CSA_OldestTxSet in the CSA frame just        sent were different, start the RetransmitTimer. If the timer was        previously running, then cancel it and restart it.    -   9. If one or more status flags have been deleted, then recompute        the network operating mode and/or priority mapping function due        to changed status flags. The mode/mapping recomputation is        performed if DeleteSet, as computed below, is not empty (0):        DeleteSet=OldestTxSet & ˜(PrevTxSet|CurrentRxSet)        If the RetransmitTimer expires, a current CSA frame is sent for        this station with the CSA_Opcode set to 0 (Announce). The timer        is not restarted. The CSA protocol does not itself perform        network mode selection, but simply provides a distribution        mechanism for configuration flags.

There is a cost of slightly lower maximum attainable bandwidthassociated with lower PHY priorities in the HPNA V2 MAC protocol if adefault mapping scheme of link layer to PHY layer priorities isemployed. This cost becomes especially burdensome when onlylower-priority traffic is being carried on the network. Therefore, theCSA protocol includes procedures for remapping lower LL priorities tohigher PHY layer priorities when no station on the network is sendingtraffic marked for those higher priorities. The choice of Physical Layer(PHY) priority for a given frame is based on its assigned Link Layer(LL) priority. The default mapping from LL priority to PHY priority isspecified below. The LL priority of a frame at the sender is conveyed tothe receiving station in order to allow proper recovery of link layerprotocol at the receiver. This requires either a fixed, one-to-one,mapping of LL to PHY priorities, or some mechanism for carrying the LLpriority within each frame. The LARQ protocol, defined below, carriesthe assigned LL priority from a sending station to a receiving station,providing the required mechanism, and thereby creating the opportunityto apply non-default LL to PHY priority mappings, which in turn, allowsfor higher maximum attainable bandwidth. A station may optionally use an802.1q header to convey the LL priority. However, since support for802.1q headers is optional, a station employing this method attempt todetermine that all receivers of the frame support the use of 802.1qheaders. Stations that do not support 802.1q headers are unlikely toproperly receive frames that include an 802.1q header. When theassignment of a Physical layer priority to the frame occurs, any changesto the PHY priority remapping function due to the use of a new priorityshould already have been made. The driver uses the remapped PHY priorityto transmit the frame (including placing this value in the Frame ControlHeader) unless the frame has no LARQ header, in which case the defaultLL-to-PHY mapping is used. The LL priority of received frames indicatedup the protocol stack by the driver (before any reassignment due to aLARQ or 802.1q header) is determined using the default PHY-to-LLpriority map, except that Minimal Profile stations indicate the priorityof all frames as LL 0. The mechanism that guarantees correct LL priorityfor received frames is the restoration of LL priority from the LARQ (oroptionally, 802.1q) header. LARQ header processing is always performedafter the default LL priority has been assigned in the receive path. TheIEEE 802.1p characteristics places the default (unassigned/best-effort)priority above both priorities 1 and 2, when an 8-level priority systemis in use. Therefore, Link Layer priority 0 will be mapped above both LL1 and LL 2 for default Physical Layer priority assignment. IEEE 802.1pdesignates priority level 7 for Network Control and priority level 6 fortraffic requiring latency of <10 msec (typically characterized asvoice-like traffic). However, on HPNA V2 networks, PHY priority level 7is reserved for traffic requiring latency of <10 msec, and NetworkControl traffic is redirected to HPNA PHY priority level 6. So thedefault mapping for LL to PHY priorities includes the swapping ofpriorities 6 and 7. For transmitted frames, the set of LL priorities [0,1, 2, 3, 4, 5, 6, 7] are by default mapped in order to the following setof PHY priorities [2, 0, 1, 3, 4, 5, 7, 6]. For received frames, PHYpriorities [0, 1, 2, 3, 4, 5, 6, 7] are, by default, mapped to LLpriorities [1, 2, 0, 3, 4, 5, 7, 6]. The PHY priority remapping isperformed below LARQ in the protocol stack, and is not applied to thepriority field in the LARQ (or optionally, 802.1q) header. PHY priorityremapping is not performed on data frames (those that are not linkcontrol frames) unless a LARQ (or optionally, 802.1q) header has beenadded with the original LL priority. PHY priority remapping is performedon Link Control Frames. Without priority mapping, a station would passthe original LL priority into-the driver, where that value would be usedto select the associated PHY priority from the default map. Withpriority remapping, the default-assigned PHY priorities are increased tomake use of higher PHY priorities that would otherwise be unused. Theremapping function is simple. For each PHY priority P that correspondsto an in-use LL priority, the new priority P′ to use is that priorityincreased by the number of higher unused priorities. For example, if [1,3, 4, 7] are in use, then priority 4 will be increased by 2 to 6, sincethere are two higher unused priorities (5,6). The tables shown in FIGS.51 a and 51 b contain a few more examples, including the defaultLL-to-PHY translation. The columns in the tables represent LL prioritiesbefore mapping. The left hand section shows some sets of in-usepriorities, with the right-hand section showing the new PHY prioritythat the driver should use in each case. The cross-hatched entries showmappings that no sender is using. However, if there is any possibilityof an implementation sending with an out-of-date mapping, or sending apriority that hasn't been included in the mapping, then it always usesthe priority of the next lower valid mapping. Consider the followingexample. If the CurrentInuse, are [0, 1, 4, 7], then the correspondingset of in-use PHY priorities is [2, 0, 4, 6]. Then increase each by thenumber of missing higher priorities:, 2->5, 0->4, 4->6 and 6->7. Just tobe safe, the any unused PHY priorities are also remapped to the newvalue of the next lower in-use priority, giving: 1->4, 3->5, 5->6, 7->7.So the in-use LL priorities [0, 1, 4, 7] result in transmitting PHYprioritie s [5, 4, 6, 7]. A complete map for all the LL priorities addsthe remaining remapped values for the default priorities correspondingto the unused LL priorities: LL[0, 1, 2, 3, 4, 5, 6, 7] gives PHY[5, 4,4, 5, 6, 7, 7].

Now turning to the Limited Automatic Repeat reQuest (LARQ) in moredetail, the operation of which is set forth in pending U.S. patentapplication Ser. No. 09/316,541 entitled “Limited Automatic RepeatRequest Protocol For Frame-Based Communications Channels” which isincorporated herein by reference. This is a protocol that reduces theeffective error rate when frame errors occur. Its primary distinctionfrom similar, sequence number-based protocols is that it does notguarantee reliable delivery of every frame, but instead conceals errorsin the physical layer through fast retransmission of frames. The goal isto significantly enhance the usability of networks that may, at leastoccasionally, have frame error rates (FER) of 1 in 10⁻² or worse.Protocols such as TCP are known to perform poorly when FER gets highenough, and other applications, such as multi-media over streamingtransport layers, are also susceptible to poor performance due to highFER conditions. The protocol provides a negative acknowledgment (NACK)mechanism for receivers to request the retransmission of frames thatwere missed or received with errors. There is no positive acknowledgmentmechanism. There is no explicit connection setup or tear-down mechanism.A reminder mechanism gives receivers a second chance to detect missingframes when relatively long gaps (in time) occur between frames. LARQfunctions as an adaptation layer between the Ethernet link layer (layer2) and the IP network layer (layer 3). It is commonly implemented in thedevice driver. Stations implement LARQ per “LARQ channel”, where a LARQChannel is identified by the tuple {source address, destination address,priority}. Stations may enable or disable LARQ processing on a channeldynamically, based on information about network frame error rates.However, in a preferred embodiment it is recommended that LARQ be leftenabled at all times, since the per-packet processing overhead is quitelow, and the complexity associated with enabling and disabling theprotocol (including determination of appropriate parameters) probablyoutweighs any likely performance gains. Stations should implement LARQ,and if they do so, they use the specified control frame formats and usethe recommended procedures defined below. Stations not adding LARQ (oroptionally, 802.1q) headers do not remap PHY priorities, and treat allreceived traffic as “best effort”, that is, all traffic is assigned toLink Layer Priority 0. Stations may choose to add LARQ headers ontransmitted frames with the LARQ_NoRtx flag set to 1. This flagindicates that the station does not retransmit frames for this channel,but adding the LARQ header allows the station to use PHY priorityremapping since the LL priority of successfully received frames will berestored from the LARQ header. All stations are capable of removing LARQheaders from received frames (de-encapsulating the original payloads).Furthermore, if the implementation supports multiple LL priorities inits receive protocol processing, then it restores the LL priority fromthe LARQ header, if one is present. If a station does not implementLARQ, then it drops LARQ control frames and it discards frames marked asretransmissions in the LARQ header. The text below uses the terms“insert” and “remove” when discussing LARQ headers. The formaldefinition of the LARQ frame format provides a Next Ethertype field thatcontains the original frame's Ethertype value. In practice, it willgenerally be the case that LARQ frames will be created by inserting the8 octets starting with the Ethertype 0x886c into the original framebetween the Ethernet header's source address and the original frame'sEthertype. The original frame's Ethertype becomes relabeled as the NextEthertype field of the final frame. The LARQ header carries LLC priorityacross the network. The use of 802.1q headers is not required for thisfunction, and V2 drivers are not required to support the use of 802.1qheaders for conveying priority. FIGS. 52 a, 52 b, 52 c, 52 d, 52 e and52 f. 1–52 f. 2, depict the LARQ Reminder Control Frame, the LARQ NACKControl Frame, the LARQ Encapsulation Frame, the LARQ_EncapsulationHeader Data, the LAEQ Control Header Data, and various terms anddefinitions, respectively. LARQ is defined for operation on simplexlogical channels. A separate logical channel is defined for eachcombination of Ethernet destination address, Ethernet source address andlink layer priority. There is no explicit channel setup procedure. A newchannel is implicitly defined when a station chooses to send LARQencapsulated frames for a new combination of DA, SA and link layerpriority. The station that sends such frames (usually the owner of theSA, except in the case of a bridge masquerading as SA) is the sender forthe channel. Each channel has a single sender. Any station that receivesthe frames and processes the LARQ headers is a receiver. There may beany number of receivers. Receivers operate independently. Variables andParameters of the sender operation are set forth in FIG. 53. For asender (new channel) implementation-dependent parameters are selected,if necessary, and an initial value for Send Sequence Number is selected;The link layer priority for the frame is determined in animplementation-dependent manner, for instance, by examining the 802.1 ppriority passed along with packets in newer NDIS implementations. Thelogical channel state information is accessed for the DA, SA and linklayer priority of the frame. The Send Sequence Number, modulo 4096 (thesize of the sequence number space)is incremented. The LARQ header isbuilt with the new value of Send Sequence Number, and the MultipleRetransmission flag set to 0. The Priority field in the LARQ header isset to the Link Layer priority value specified for the frame. If nopriority is specified, then the priority is set to 0. A LARQ header(short form control frame format with LARQ_hdr data) is inserted betweenthe SA and the Ethertype/Length field of the original frame. The newframe is eight bytes longer than the original. A copy of the frame issaved and the frame is then sent. The reminder timer for the channel isrestarted. A save timer is started for the sequence number. When noother resource limitations apply, a sending station normally saves aframe for Maximum Save Interval, which corresponds to Maximum HoldInterval used by LARQ receivers. For the send to process a NACK ControlFrame the priority and Original Destination Address (NACK_DA) are readfrom the LARQ NACK header. The logical channel state information for theSender channel is accessed, where the channel DA is the NACK_DA and thechannel SA is the Ethernet DA from the Nack control frame. The NACKCount in the LARQ header indicates the number of sequence numbersrequested for retransmission. The first indicated sequence number is thevalue Sequence Number in the NACK header, followed by the next (NACKCount−1) sequence numbers. For each indicated sequence number startingwith the first:

-   -   If a copy of the original frame is no longer available, go to        the next sequence number.    -   If the most recent retransmission of the frame is within Minimum        Retransmission Interval of the current time, go to the next        sequence number.    -   Prepare a copy of the original frame with its original LARQ        header for retransmission.    -   Copy the value of the Multiple Retransmission Flag from the NACK        header into the LARQ header of the frame to be retransmitted.    -   Set the LARQ_Rtx flag to 1.    -   Send the retransmitted frame.        A retransmission is not sent if a received Nack control frame        has an error. If the reminder timer expires, a Reminder control        frame is created, with the Sequence Number set to the current        value of Send Sequence Number for the channel. The priority for        the Reminder control frame is the same as the priority for the        channel. The frame is then sent and the reminder timer is not        restarted for the channel. The save timer sets an upper bound on        how long frames will be saved by a sender for possible        retransmission. If set too long, host resources may be wasted        saving frames that will never be retransmitted. This timer is        conceptually implemented per sequence number. Any resources        associated with the saved frame are released. A LARQ        implementation requires careful attention to resource        management. The resources include the buffers used for saving        copies of data for retransmission, the buffers and other        resources used to manage the re-ordering of frames to        incorporate retransmissions, and the various timers used to        govern proper behavior and efficient protocol operation. Saved        copies of frames are kept for Maximum Save Interval (default is        150 ms), other considerations notwithstanding. The maximum        number of saved frames for any channel, are a function of the        maximum rate that new frames may be generated. Very slow devices        might usefully save only a couple of frames for retransmission.        A high-speed device serving video streams might save 100 or more        frames for a single channel. Senders that save relatively few        frames are more likely to receive NACK control frames for        sequence numbers that can no longer be retransmitted. Such        behavior is inefficient, but causes no other problems. The        description of correct protocol for receiver operation channel        variables and parameters are set forth in FIG. 54. The actual        implementation may vary so long as the behavior remains        unchanged. When a data frame with a LARQ header or a LARQ        Reminder control frame is received with a new combination of DA,        SA and link layer priority, the receiver initializes state        information for a new channel. The primary piece of state        information is the Current Sequence Number for the channel.        Current Sequence Number is initialized to the sequence number        immediately preceding that found in the LARQ header of the        received frame. This assignment takes place prior to processing        the received frame and results in the frame either appearing to        be the next expected data frame, or the reminder for the next        expected data frame. With regard to receiver LARQ data or        reminder frame, the channel state information is looked up based        on the Ethernet DA and SA in the received frame plus the Link        Layer priority from the LARQ header. A new channel is setup if        necessary. If the received sequence number of the received frame        is out of sequence, the channel state may be reset. If the        sequence number (before resetting) is old, and the Forget timer        has expired, then the sequence space may be reset to the value        of the received frame's sequence number. If the received        sequence number is newer than the Current Sequence Number (after        any reset of the sequence number space) then new sequence number        processing steps are performed as set forth below, otherwise the        old sequence number processing steps is performed. With regard        to receiver LARQ frames with CRC or other errors, for best        performance, implementations allow the LARQ protocol module to        process errored frames, such as those with payload CRC errors.        This will, in many cases, allow Nack indications to be sent more        quickly since the receiver will not have to wait for the next        frame to detect the loss. At the same time, it provides a second        opportunity for detecting lost frames at the end of a sequence,        when a later Reminder would be the only protection. If errored        frames are used, they are used only to detect a very small of        missing sequence numbers for an existing channel (one missed        frame is recommended). In particular, if the frame appears to        have a valid LARQ header, and the frame's source MAC address,        destination MAC address, and LARQ header priority match an        existing logical channel, and if the sequence number is (Current        Sequence Number+1), then this frame is treated as a Reminder        control frame for the purposes of processing. Note that Reminder        control frames are always dropped after processing. In all other        cases, the errored frame is dropped with no further processing.        A new channel is not set up if the frame has an error. A        retransmission is not sent if a Nack control frame has an error.        A channel is not reset (for sequence numbering purposes) for an        errored frame.

With regard to the receiver involving a new sequence number, if theframe has an error indicated by a lower layer driver, such as a CRCerror, and the sequence number of the frame is anything other than(Current Sequence Number+1), then the frame is dropped with no furtherprocessing. Otherwise, the frame is processed as a Reminder controlframe. If the difference between the new sequence number of the receivedframe and the oldest missing sequence number is greater than (MaximumReceive Limit−1), the following steps are repeated until the acceptablelimit is reached.

-   -   Cancel the Nack retransmission timer and the lost frame timer        for the oldest missing sequence number.    -   If there is a saved frame for the next sequence number, then        deliver in-sequence frames to the next layer above until the        next sequence number with a missing frame is reached (which may        be the next expected sequence number for the channel,    -   (Current Sequence Number+1)). The value from the Priority field        from the LARQ header for each frame is delivered to the next        layer along with each associated frame.        If the sequence number is the next expected sequence number        (Current Sequence Number+1) and the frame is a good data frame        and there are no older missing sequence numbers, then the frame        is sent up to the next layer. If the sequence number is newer        than (Current Sequence Number+1), or is a reminder for (Current        Sequence Number+1), then one or more Nack control frames is sent        requesting retransmission of the missing frame(s). The        destination address for the Nack is the source address of the        received frame. The source address is this station's MAC        address. The destination address of the received frame is placed        in the original destination address field (NACK_DA) in the LARQ        Nack control frame header. The Multiple Retransmission flag is        set to 0. The [first] missing sequence number is placed in the        sequence number field. The priority for the Reminder control        frame is the same as the priority for the channel. If multiple        Nack control frames are sent, the earliest sequence number is        sent first. For each missing sequence number a Nack        retransmission timer is started, set to expire at the current        time plus Nack Retransmission Interval. For each missing        sequence number, a lost frame timer is started, set to expire at        the current time plus Maximum Hold Interval. If the frame is a        good data frame and was not delivered to the next layer it is        saved. If the frame is a reminder frame (or an errored data        frame) it is dropped. The Current Sequence Number is then        advanced to the sequence number in the received frame.

With regard to the receiver regarding an old sequence number, if thesequence number is the same or older than Current Sequence Number, thenit will generate no control frames, although it may itself be dropped,held, or sent up to the next higher layer, possibly causing other heldframes to be sent up as well. It may cause the cancellation of a Nackretransmission timer or lost frame timer associated with that sequencenumber. If the frame is not a good (e.g. bad CRC) data frame, or it'ssequence number is older than the oldest missing frame, or it hasalready been received (this is a duplicate retransmission), or it is aReminder frame, then the frame is dropped and further processing isskipped for this frame. The Nack retransmission timer and the lost frametimer is cancelled for the sequence number. If the sequence is not theoldest missing sequence number, then the frame is saved. If the sequencenumber is the oldest missing sequence number, then the frame isdelivered up to the next higher layer. If there is a saved frame for thenext sequence number, then in-sequence frames are delivered to the layerabove until the next sequence number with a missing frame is reached(which may be the next expected sequence number for the channel). Thevalue from the Priority field from the LARQ header for each frame isdelivered to the next layer along with each associated frame.

With regard to the Receiver and Nack retransmission timer expiration, ifa Nack retransmission timer expires, then another Nack control frame issent for the associated sequence number. The priority for the Nackcontrol frame is the same as the priority for the channel. Multiplesequence numbers may be nacked at the same time, if their timers expireat similar times. The Multiple Retransmission flag is set to 1 for Nackcontrol frames sent as a result of retransmission timer expiration.While there is no explicit limit on the number of Nack control framessent for a particular sequence number, it should be noted that the Nacktimer is canceled if the frame is received or if the sequence number isdeclared lost.

With regard to the Receiver and lost frame timer expiration, the lostframe timer is implementation dependent. Its purpose is to set an upperbound on how long frames will be held before they are sent up when aframe is really lost. If set too long, network resources may be wastedon NACK control frames sent for frames that the sender on the channelwill never retransmit. Further, higher layer transport timers may alsobecome involved. The default value of 150 ms is strongly suggested as anupper bound. Upon expiration, the sequence number is declared lost,resulting in the cancellation of the Nack retransmission timer and thelost frame timer for the sequence number. If there is a saved frame forthe next sequence number, then send up in-sequence frames until the nextsequence number with a missing frame is reached (which may be the nextexpected sequence number for the channel). If the lost frame timers formultiple sequence numbers expire at the same time, then the timers areprocessed in sequence from oldest to newest. With regard to theReceiver, a forget timer is provided. The forget timer is animplementation dependent mechanism to allow a receiver to reset thesequence number space of a channel when a received sequence number isnot the next expected (Current Sequence Number+1) and a relatively longinterval has expired since the last frame received on the channel. Onceexpired, a receiver will accept any unusual sequence number as the nextexpected sequence number, allowing for undetected resets of otherstations, disconnection from the network, etc. The definition of“unusual sequence number” is implementation dependent, but generallymeans any old sequence number or any new sequence number that is notclose to the current sequence number, where “close” is 1 or some othersmall integer. A one second default is suggested.

With regard to Receiver resource management, in general, the receiverwill want to set upper bounds on the number held frames per channel andthe number of held frames across channels. The bounds may vary based onthe priority of the channel. Timer intervals may vary based on factorssuch as the priority of the channel, or measured intervals forsuccessful retransmissions. The description above suggests per-sequencenumber timers. This is for descriptive purposes only, and does not implyany implementation mechanism.

It should be noted that with regard to the Link Layer Protocol, thereare certain vendor specific formats. Referring to FIGS. 55 a and 55 brespectively, two types allow vendor-specific extensions which may bereasonably handled by implementations that do not otherwise supportthem. The vendor specific short frame format set forth in FIG. 55 aallows short control messages and encapsulation headers, while thevendor specific long frame format set forth in FIG. 55 b allows otherextensions that require longer messages.

With regard to Minimal Link Protocol Support Profile for HPNA V2 LinkProtocols, the Minimal Link Protocol Support Profile for HPNA V2 LinkProtocols allows less complex implementations of the HPNA V2characteristics. While each of the component protocols serves animportant function in the operation of the network, it is possible toimplement minimal support for some of, the more complex protocols thatis compatible with fully functional implementations and does not detractfrom the overall performance of other stations. The shorter name,Minimal Profile, will be used in the following description. Thealternative is full support of all the link protocols, called the FullLink Protocol Support Profile, or Full Profile for short. A MinimalProfile station can send only best effort data traffic, and treats allreceived traffic as best effort. A Minimal Profile station cannotadvertise or use optional features that may be defined in the future.Due to the lack of support for LARQ, a Minimal Profile station may seedramatically reduced network throughput. A Minimal Profile station isable to handle all HPNA V2 Link Protocol frames, which are those markedthe HPNA Ethertype 0x886c in the Ethernet header of the received frame.This includes dropping control frames with unknown subtypes andde-encapsulating data frames with unknown subtypes. The length field isused to locate the Next_Ethertype field in order to determine whetherframes are control or data (encapsulated) frames. A Minimal Profilestation implements the standard HPNA V2 Link Integrity function,including suppression of LICFs in native V2 mode. A Minimal Profilestation implements the full set of rate-selection functions required foroperation in both 1m2 mode and V2 mode using the 2 MBaud band. A MinimalProfile station properly handles frames with LARQ headers. It dropsreceived control frames. It properly removes LARQ headers from dataframes. In addition, if the LARQ header on a data frame has theretransmission flag set, then the frame is dropped in order to preventduplicate and out-of-order frames. A Minimal Profile station adds LARQheaders to data frames being transmitted, setting the priority to 0 andthe LARQ_NORTX flag to 1 in the LARQ headers. If LARQ headers are added,then the minimal station may use priority remapping based on prioritystatus information received in CSA messages. Alternatively, if LARQheaders are added, the minimal station may use default priority mapping.A Minimal Profile station listens to CSA Control Frames and performsmode selection based on the configuration flags received(ConfigV1,ConfigCompatibility,ConfigV2). In particular, it uses theunion of the CSA_CurrentTxSet and CSA_CurrentRxSet as the set of in-useflags. A Minimal Profile station does not send CSA Control Frames, andcan therefore never advertise optional features, or use non-defaultpriorities. In addition to control frames, a Minimal Profile stationonly sends normal data frames using the default priority assigned tobest-effort/unspecified QOS. The Link Layer priority value for this QOSis 0. If the station is not adding LARQ headers, then data frames issent using the default physical layer priority for link layer priority0. (i.e. it uses physical layer priority 2.) If LARQ headers are beingadded as specified above, then the LARQ header priority field is set to0, and the station again uses the default remapping function for linklayer priorities to determine the actual Physical layer priority to usefor Link Layer priority 0. (i.e. it uses physical layer priority 2.) AMinimal Profile station only indicates LL priority 0, if any priority isindicated, for received frames, regardless of the physical layerpriority or priority value in a LARQ header. In support of MinimalProfile stations, a minor addition is also needed for the CSAcharacteristics for Full Profile stations. Any station that is notsending CSA frames, but which is determined to be a V2 station as aresult of traffic received from that station, is treated as if itadvertised a default set of status flags, including no supportedoptions, only LL priority 0 in use, and highest supported station typeV2.

Homenetworking Further Implementation Details

Certain further aspects of the embodiments of the present invention asdescribed in more detail below. These aspects include: carrier sense forseverely distorted networks, collision detector for severly distortednetworks, scrambler and descrambler initialization circuits, gainestimation circuit for burst modem, rate negotiation and rate selectionalgorithms, Split Winding Transformer for Modem Transceiver S/NOptimization, and Transmit Off Switch for Modem Receiver Noise Reduction

Carrier Sense for Severely Distorted Networks

Now turning to the carrier sense function in more detail, a preferredcarrier sensing embodiment which is particularly useful forseverely-distorted networks is described. On a typical Ethernet bus, alltaps are terminated in the characteristic impedance of the line tominimize reflected signal power. Because reflections are insignificantand the signal-to-noise ratio (SNR) at each receiver is very high, asimple carrier sense technique (e.g. level detector with a fixedthreshold) may be used to determine when the medium is busy. Inresidential networking over pre-installed wiring (e.g. phone wiring,power wiring), attenuation may be high due to wall jacks and unused wiresegments that are not terminated with the characteristic impedance ofthe wire. There will also be severe reflections for the same reason. Thereceiver SNRs may be low (10 dB or lower in some cases). In addition,the problem is complicated by the fact that every path between twostations on the network has a different channel impulse response. On onepath, two stations may communicate at a high rate (e.g. 8 bits/symbol),while all other paths only support 2 bits/symbol. The implication ofthis example is that the demodulator may not be used as the method ofcarrier sense in such a network, as all stations on the network are ableto delineate frames, even those whose payloads may not be demodulateddue to insufficient SNR. Beyond even these complications, there isimpulse noise, which may result in false carrier detection with certaintypes of detectors. In accordance with the present invention a-detectoris provided for precisely determining the start of a frame (within 1microsecond) in a severely-impaired CSMA/CD network. In addition, thisdetector determines the start of a frame with sufficient precision togenerate a channel model with a small number of adjustable coefficientsfor generating decision-feedback equalizer weights. In accordnace withthe present invention, a preamble format is provided in which Midentical copies of the same k*n-symbol quadrature phase-shift keying(QPSK) sequence are transmitted sequentially. This k*n-symbol sequenceis spectrally white over an k*n-symbol span (has a single non-zerocircular autocorrelation value). Further, the k*n-symbol QPSK sequenceconsists of k sequentially-transmitted copies of an n-symbol subsequencethat is spectrally white over an n-symbol span. Furthers a detector forprecisely determining the end of a frame (within a 4-microsecond window)in a severely-impaired CSMA/CD network is provided. In accordance withthe present invention an n-symbol sequence that is spectrally white overan n-symbol span that delimits the end of a burst and enables thisdetector is provided. By keeping the end-of-frame detection uncertaintylow, the efficiency of the network is increased.

The carrier sensor, for example carrier sense 1100 of FIG. 30, consistsof two components: one which detects the start of frame and one whichdetects the end of frame. The carrier sense circuit takes an input fromthe medium access controller (MAC), which forces reset. The decisionlogic depends on the state information, per the table set forth in FIG.56.

With regard to the start-of-preamble detector, which is described belowin conjunction with FIG. 59, complex samples at L times the nominaltransmitted symbol rate are the input to this detector. Typically, Lwill be 2, 4, 8, or 16. The complex samples are generated by a filterwhich performs nearly a Hilbert transform on its input. In addition, theinput samples are band-pass and notch filtered to attenuate noise andinterference while minimally reducing the channel capacity. The circuitembodiment in accordance with the present invention consists of aspecial filter whose coefficients are matched to the preamble symbolsequence and a detector which performs near-optimal detection of thepreamble in the presence of additive white Gaussian noise. Thefilter/detector is not a true matched filter detector in the textbooksense, because: (1)the filter is not matched to the input samplesequence but, rather, the input symbol sequence, and (2)the filterincludes additional delay elements to minimize the probability of afalse trigger immediately before the correct start of burst position.There are two possibilities for start of preamble detectors: a simple,low-delay detector and a more complex, robust detector. Both areincluded in the description of the embodiments below. The circuit inaccordance with the present invention is designed to operate in networksin which the insertion loss between any two points is less than 38 dB.Since the carrier sense function must operate before the gain can beadjusted, the minimum SNR at which the system must operate is calculatedas:SNR>PSNR(10-bit ADC)−PAR(4-QAM preamble)−L _(insert)PSNR is the peak signal to noise+distortion for the analog-to-digitalconverter (ADC), which is about 60 dB. The worst-case PAR for thepreamble is about 10 dB. Note that if the line noise floor is greaterthan the ADC noise+distortion floor, the maximum tolerated insertionloss will be less than 38 dB. So, the start of preamble detectionfunction must operate reliably down to about 12 dB SNR. Reliableoperation is defined as no more than one missed detection in 10⁵ actualframes and no more than one false alarm in 10 seconds in additive whiteGaussian noise (no valid frames). Missed detection performance shouldimprove with increasing SNR. Reliable detection tends to require longerfilters and more averaging. Unfortunately, increasing reliability hasthe side-effect of increasing the medium access slot times. Because ofthe need to minimize the slot time, the start of preamble detector mayconsist of two matched filter detectors. One is a “first-pass”, shortmatched filter detector which is used for determining slot boundaries(to minimize the slot duration). The second uses a matched filter whichspans one entire copy of the training preamble, for reliable detection.The first-pass detector produces a “transmit hold-off” signal, which isused only to inhibit transmission until the second-pass (longer filter)detector makes a more reliable determination of medium state. Thesecond-pass start of preamble detector uses a matched filter withaveraging and an average power estimate to determine the start of framewithin +/−1-microsecond intervals. Therefore, in accordance with thepresent invention, both a circuit in which the low-delay and robustdetectors are used in conjunction and also a circuit in which only therobust detector is used is provided.

With regard to the low-delay detector, it uses a filter matched to thefirst n symbols of the preamble. The filter coefficients are the first nsymbols of the preamble in reverse order, complex-conjugated, theninterspersed with L zeros per symbol. If the first n symbols are [s₀,s₁, . . . s_((n−1))] then the filter coefficients are [s_((n−1))*, 0, 0,0, s_((n−2))*, 0, 0, 0, . . . s₀*, 0, 0, 0], when L=4. “*” indicatescomplex conjugation of the symbol value. The bit widths, shown in FIG.57 as r, r+1,q, etc., are merely examples in one particular embodiment,and the invention is not limited to any particular datapath widths. “j”is the sample (time) index in FIG. 57. Note that, because the preambleconsists of only QPSK symbols, no multiplications (only additions andsubtractions) are required. The output of the MA block is computed asmax (x_(i), x_(q))+½^ min (x_(i), x_(q)), where x_(i) is the in-phasecomponent of the complex sample and x_(q) is the quadrature-component,with rounding. The output of the matched filter in this one embodimentsaturates at r+1 bits twos-complement, but other outputs are possiblewithin the scope of this invention. AVG may be either a simpleL*n-sample moving average or a one-pole smoothing filter withalpha=1/(L*n).

Referring briefly to FIG. 58, which is described in more detail below, acircuit block diagram is shown giving an example of an averaging circuitfor L*n=16 samples. The (roughly) equivalent moving average would sumthe 16 samples, then shift the result right by 4 bits with rounding.Therefore, in accordance with the present invention a circuit isprovided that enables rapid detection of the start of a burst using thefirst n symbols of the preamble and no multiplication operations. Thisallows delineation of medium access slot boundaries.

With regard to the robust detector, it uses a filter matched to thefirst k*n symbols of the preamble, described above. The filtercoefficients are the first k*n symbols of the preamble in reverse order,complex-conjugated, then interspersed with L zeros per symbol. If thefirst k*n symbols are [s₀, s₁, . . . s_((k*n−1))], then the filtercoefficients are [s_((k*n−1))*, 0, 0, 0, s_((k*n−2))*, 0, 0, 0, . . .s₀*, 0, 0, 0], when L=4. “*” indicates complex conjugation of the symbolvalue. The bit widths, shown in the FIG. 59 as r, r+1,q, etc., aremerely examples in one particular embodiment, and the invention is notlimited to any particular datapath widths. Similarly, the thresholdsdepicted in the figure are considered “good values”, but are not arequirement of the invention. The thresholds are adjustable. The delaysbetween the delays between the h_(match) path and the power estimationpath, D^(v1) and D^(v2), are used to account for the differences ingroup delay between the paths. If these delays are not included, theprobability of a false trigger slightly before the beginning of a framemay be increased. Note that, because the preamble consists of only QPSKsymbols, no multiplications (only additions and subtractions) arerequired. The output of the MA block is computed as max(x_(i), x₉)+½*min(x_(i), x_(q)), where x_(i) is the in-phase component of the complexsample and x_(q) is the quadrature component, with rounding. The outputof the matched filter in this one embodiment saturates at r−1 bitstwos-complement, but other outputs are possible within the scope of thisinvention. AVG1 may be either a simple L*k*n-sample moving average or aone-pole smoothing filter with alpha=1/(L*k*n) and a variable outputscaler. AVG2 may either be a L*n-sample moving average or a one-polesmoothing filter with alpha=1/(L*n) and a variable output scaler. Noteagain that the “matched filter” in this detector is not a true “matchedfilter”, since it is not matched to the expected input sample sequence:there is no a priori knowledge of the wire network's impulse response.Therefore, in accordance with the present invention a circuit isprovided that enables very robust detection of the start of a burstusing the first k*n symbols of the preamble and no multiplicationoperations. This enables efficient channel estimation (fewercoefficients), reliable detection of the start of a burst in SNRs as lowas 3 dB, and accurate automatic gain control.

Referring back to FIG. 59, specific operational aspects of the start offrame detection are described in more detail. One aspect is the start offrame detection. In upper portion 3010 the first stages of carriersense/start of preamble detection is shown. In lower portion 3012 theremaining stages are shown. Accordingly, the carrier sense processingstarts at the upper left portion of FIG. 59 and ends at the lower rightportion of FIG. 59. Input 3014 has r bits, which in a preferredembodiment is at 8 Msamples/sec. Matching filter/correlator 3016receives the r bits, and filters the input using filter coefficientswhich are a time-reversed sequence copy of the preamble sequence. Theoutput of filter 3016 is provided to magnitude approximator 3018 andsquaring function 3020. Magnitude approximator 3018 provides a realoutput for which on one squaring operation is needed, avoiding the needfor a multiplier function. The output of squaring function 3020 is inputto low-pass filter 3022. Low pass filter 3022 smoothes the input theretoand provides output Z_(j). At input 3014 r is also fed into an energydetection computation where magnitude approximation 3024 is performed,then a squaring operation 3026, then a longer duration low-passfiltering 3028, and then performing a low-pass filtering operation 3030comparable to that of low-pass filter 3022, providing an output zh_(j).z_(j) is then put through logarithm function 3032 to allow measuring ofratios avoiding division operations. zh_(j) is similarly put throughlogarithm function 3034. The output from logarithm function 3032. Twotests are performed during the carrier sense computation at comparefunctions 3036 and 3038. Where inputs A and B respectively are comparedbased upon a threshold, e.g., 9 dB threshold input into compare function3036 and 3 dB threshold input into compare function 3038. In other wordsa calculation is performed to determine if A−B is>than the threshold.Delays 3040, 3042, 3044 are provided between the logarithm functions andthe compare functions. In essence, with regard to the Z_(j) processingof portion 3012, the smoothed low-pass filtered output of the matchedfilter is compared with a delayed copy of itself, as provided by delay3040. In addition delay function 3044 is applied to the output oflogarithm function 3032 providing a slightly delayed input to comparefunction 3038. With regard to the zhj processing, delay function 3042and maximizing function 3046 is applied to the output of logarithmfunction 3034 to provide a sampled maximum to avoid getting a falsetrigger. Therefore compare function 3038 compares that smoothed low-passfiltered output is greater than the Z_(hj) output of the energydetector. The output of comparator functions 3036 and 3038 is thenprovided to AND function 3040, a match if both inputs are true and a nomatch if one is not true.

Referring now to FIG. 58 in more detail, low-pass filtering function3022 (and its counterparter function 3030) is depicted in more detailand is considered an implementation of an Infinite Impulse Response(IIR) filter. Left by 4 bit shift 3042 is applied to input signals N,the output of which is applied to adder 3044. The output of adder 3044is provided to round to the nearest integer function 3046 which providesa signal which is sent to right by 4 bit shift 3048, then to one clockdelay 3050. The output of one clock delay 3050 is provided to left by 4bit shift 3052, the output of which is applied to subtractor 3054 alongwith an output of delay 3050. The value of the recursive path is thenset thereby. Rounder 3056 takes the N+8 signal from adder 3044 andprovides its output to Right by 8 bit shift 3058 to provide N bitsoutput. On the other hand filter 3028 is a moving average filter whichtakes N inputs sums them together and divides the sum by N.

With regard to the end-of-frame detector, end of frame detection iscomplicated by the need to avoid prematurely detecting the end of aburst and the possibibility that there can be: (1) a long run ofinnermost constellation points in a large transmitted constellation—inthis case, the receiver attempting to determine the end of frame may beunable to demodulate the signal because the SNR is not sufficient (theinnermost points are lost in noise);and (2) a long run of the same, ornearly the same, constellation point—in this case, a channel with a nullin exactly the wrong place can substantially attenuate this symbolsequence. The end of frame detector uses a short filter matched to thelast n symbols of the frame (the end-of-frame delimiter), above. Again,the input is complex at L times the nominal transmitted symbol rate. Thematched filter in this case consists of L*n coefficients, which are then “end-of-frame” marker symbols, complex-conjugated, time-reversed, andupsampled by L with zero-filling. The input is applied to the matchedfilter, and the output of the filter is passed into a magnitudeapproximation circuit. The output of the magnitude approximation circuitis squared and applied to the same one-pole low-pass filter or movingaverage filter described in the sections above. The averaged output isthen applied to the same approximate 10*log 10(.) function. This outputcan be called z(j), where j is the sample index. The first criterion fordetecting the end of a frame is:z(j)−z(j−L*p)<thd_offA reasonable value for thd_off in one embodiment is 8 dB. A reasonablevalue for L*p in microseconds is 12. Therefore, in accordance with thepresent invention other values of thd_off and L*p within scope the scopeof the present invention. If this test passes, the z(j−L*p) value isstored, and the same test is applied on the L*k*n subsequent samples,replacing z(j−L*p) with this stored value. If k1 of these tests pass,the end of frame is declared. The end-of-carrier detector describedherein is capable of determining the end of a received frame within a+/−2 microseconds interval, and can be considered as a circuit thatperforms two tests in order. Therefore, in accordance with the presentinvention a circuit is provided that enables rapid detection of the endof carrier using the last n symbols of the burst and no multiplicationoperations. This enables efficient use of the medium by keeping mediumaccess slots short.

Referring to FIG. 60, with regard to the first test of theend-of-carrier detector, it uses an n-symbol matched filter, almostexactly as the first-pass carrier sense. However, note that, while thecoefficients are identical, the delay between compared matched filteroutputs is not 2*n−1 symbols, but p>(2*n−1) symbols. This is required,as the channel postcursor ISI typically far exceeds the duration of theprecursor ISI on phoneline, powerline, or wireless networks. Again, asbefore, the datapath widths are given strictly to show one particularembodiment. The exact values in the diagram do not limit the scope ofthe current invention. AVG is either an L*n-sample moving average or aone-pole filter with alpha=1/(L*n) and a variable output scaler.

With regard to the second test, when the output of the first test goesactive, the delayed value r_(j.L*p) is stored. On the L*k*n subsequentsamples, the stored value is compared with the the non-delayed r_(j)value. If the thd_off threshold is exceeded on k1 of these samples, theend of the frame is declared. One particular embodiment requires thatthe thd_off threshold be exceeded on all L*k*n samples. The statediagram shown in FIG. 61 depicts such a case.

In accordance with the present invention, a frame may not be terminateduntil a specified point after the start of the frame. In one embodiment,this point is the end of the Ethertype field of an encapsulated Ethernetframe. The present invention includes a timer to ensure that the devicedoes not remain in the BUSY state indefinitely. The present inventionfurther includes the ability to detect a third-party collision any timethe time between the second-pass start-of-preamble detection and theend-of-frame detection is less than a specified duration threshold. A“third-party collision” is one in which the detecting station was not atransmitter. An embodiment of the invention optionally can include a dBfunction implemented with two 1-entry look-up tables, a coarse 10*log10(.) table and a fine 10*log 10(.) table. The tables include unsignedvalues with m0 fractional bits. An example embodiment could be describedby the tables set forth in FIGS. 62 a and 62 b, showing coarse dB tablevalues and fine dB table values, respectively. The algorithm is:

-   1. ×=max(x, 2)-   2. Find the most significant non-zero bit in the input, x. Call the    position of this bit (0 . . . 31) b.-   3. d₁=coarse_tbl[b]-   4. if b>3, k=(x−2^(b))<<(b−5), else k=(x−2^(b))>>(5−b).-   5. d₂=fine_tbl[k]-   6. output=d₁+d₂    The output in the example embodiment yields 96 dB of dynamic range    with up to 0.25 dB resolution.

With regard to the end of carrier detector, there are two aspects whichwill be described in more detail referring again to FIGS. 60 and 61. InFIG. 60 it should be noted that the path from r input to match/no matchoutput is similar to the correlator path in FIG. 59 without the energydetection path. Since at the end of the preamble there are fourcontiguous symbols of the training preamble. Therefore, the matchedfilter is run against the entire frame looking for peaks. Towards theend of the frame there will be a small spike and dropoff. The flow inFIG. 60 provides a tentative match/no match decision, similar to that ofthe correlator path of FIG. 59. The decision is tentative to avoidpremature truncation of the frame. A state machine, as depicted in FIG.61, is provided taking in the tentative match/no match decision as shownin FIG. 60. As described above, Test 1 is match/no match. If out of Test1 the decision is yes, the state proceeds to Test 2, 0. When going fromTest 1 to Test 2, 0 the last value of peak detected is latched. Thenevery subsequent sample is compared against the latched value. If aftera number of comparisons it is determined that the signal is greater thanthe threshold value, the end of frame is determined to be detected.

Referring again to FIGS. 62 a and 62 b, the logarithm functions 3032 and3034 as described above can be implemented using the index and valuesset forth in FIGS. 62 a and 62 b.

Collision Detector for Severely—Distorted Networks

The collision detector design for networking on phone lines iscomplicated by the need to detect collisions even when the line inputimpedance changes on the time scale of a transmitted frame. Line inputimpedance changes occur with telephone hook-switch transitions, keying,and addition/deletion of devices from the network. In addition, sincethe hybrid will not exactly match the line input impedance, asubstantial amount of hybrid leakage (echo) will be present; so, it isnot possible to simply use carrier detection as a criterion for acollision when transmitting. Because the noise floor may varysubstantially over time, due to crosstalk and impulse noise, and becausea colliding signal may be attenuated by as much as 38 dB more than theecho signal, an estimate of the noise floor is needed for optimaldetection of colliding signals. This noise estimate is made eitherdirectly before the transmitted frame or during the preamble. Theclaimed invention makes this estimate during the preamble. The collisiondetector circuit in accordance with the present invention includes threefunctional aspects: (1) a channel estimator, the concepts of which aredescribed in co-pending application Ser. No. 09/585,774, entitled“Method and Apparatus for Efficient Determination of Channel Estimateand Baud Frequency Offset Estimate” and which is incorporated byreference herein; (2) a noise floor estimator; and (3) a unique fieldmatch (either SRC or SI+SRC or SRC+DST or SI+SRC+DST).

In accordance with the present invention a method and apparatus isprovided for generating a preamble sequence to facilitate channelestimation and noise floor estimation. A sequence b is defined as the 16symbols set forth below.

$b = {\begin{bmatrix}b_{0} \\b_{1} \\\vdots \\b_{15}\end{bmatrix} = \begin{bmatrix}{1 + i} \\{{- 1} - i} \\{{- 1} - i} \\{{- 1} - i} \\{1 + i} \\{1 - i} \\{1 + i} \\{{- 1} + i} \\{1 + i} \\{1 + i} \\{{- 1} - i} \\{1 + i} \\{1 + i} \\{{- 1} + i} \\{1 + i} \\{1 - i}\end{bmatrix}}$This sequence has an important property that

${\frac{1}{32}{\sum\limits_{k = 0}^{15}{b_{k}b_{{mod}{({{k + n},16})}}^{*}}}} = \left\{ \begin{matrix}{1,{n = 0}} \\{0,{n \neq 0}}\end{matrix} \right.$All symbols in this sequence belong to a 4-QAM (or QPSK) constellation.The preamble sequence is generated as four sequential copies of the16-symbol sequence b defined above. Channel estimation for the purposeof detecting collisions is performed on either the first, second andthird copies of the preamble and/or on the second, third, and fourthcopies of the preamble. Throughout the following section on channelestimation, the copies used in one estimate are referred to as thefirst, second, and third copies, respectively. The characterizationsignal is the part of the received signal used for channel estimation.This signal is defined as the second and third or third and fourthcopies of preamble in the received signal. The start of this signal atthe receiver input interface is found by simply waiting a fixed timeinterval after the start of transmission (to account for fixedpropagation delays). Referring to FIGS. 63 a–63 c, the quantitiesdescribed above are shown, assuming that the characterization signal isthe third and fourth copies of the preamble sequence.

Further, in accordance with the present invention, a method forcomputing a complex channel estimate sampled at four times the symbolfrequency of the preamble signal is provided. The complex input signalis also sampled at four times the symbol frequency of the preamblesignal. Let B represent the matrix of preamble symbol values, upsampledby four and zero-filled:

$B^{H} = \begin{bmatrix}b_{0}^{*} & 0 & 0 & 0 & b_{1}^{*} & 0 & 0 & 0 & \cdots & b_{15}^{*} & 0 & 0 & 0 \\0 & b_{0}^{*} & 0 & 0 & 0 & b_{1}^{*} & 0 & 0 & \cdots & 0 & b_{15}^{*} & 0 & 0 \\0 & 0 & b_{0}^{*} & 0 & 0 & 0 & b_{1}^{*} & 0 & \cdots & 0 & 0 & b_{15}^{*} & 0 \\0 & 0 & 0 & b_{0}^{*} & 0 & 0 & 0 & b_{1}^{*} & \cdots & 0 & 0 & 0 & b_{15}^{*} \\b_{15}^{*} & 0 & 0 & 0 & b_{0}^{*} & 0 & 0 & 0 & \cdots & b_{14}^{*} & 0 & 0 & 0 \\0 & b_{15}^{*} & 0 & 0 & 0 & b_{0}^{*} & 0 & 0 & \cdots & 0 & b_{14}^{*} & 0 & 0 \\0 & 0 & b_{15}^{*} & 0 & 0 & 0 & b_{0}^{*} & 0 & \cdots & 0 & 0 & b_{14}^{*} & 0 \\0 & 0 & 0 & b_{15}^{*} & 0 & 0 & 0 & b_{0}^{*} & \cdots & 0 & 0 & 0 & b_{14}^{*} \\\vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & ⋰ & \vdots & \vdots & \vdots & \vdots \\b_{1}^{*} & 0 & 0 & 0 & b_{2}^{*} & 0 & 0 & 0 & \cdots & b_{0}^{*} & 0 & 0 & 0 \\0 & b_{1}^{*} & 0 & 0 & 0 & b_{2}^{*} & 0 & 0 & \cdots & 0 & b_{0}^{*} & 0 & 0 \\0 & 0 & b_{1}^{*} & 0 & 0 & 0 & b_{2}^{*} & 0 & \cdots & 0 & 0 & b_{0}^{*} & 0 \\0 & 0 & 0 & b_{1}^{*} & 0 & 0 & 0 & b_{2}^{*} & \cdots & 0 & 0 & 0 & b_{0}^{*}\end{bmatrix}$here ( )^(H) represents the Hermitian transpose or conjugate transposeand where “*” indicates complex conjugation of a scalar element. Let Y₁,Y₂, and y be column vectors of received samples in the characterizationsignal:

$y_{1} = {{\begin{bmatrix}y_{0} \\y_{1} \\\vdots \\y_{62} \\y_{63}\end{bmatrix}\mspace{50mu} y_{2}} = {{\begin{bmatrix}y_{64} \\y_{65} \\\vdots \\y_{126} \\y_{127}\end{bmatrix}\mspace{31mu} y} = \begin{bmatrix}y_{1} \\y_{2}\end{bmatrix}}}$Let h be a complex 64-sample channel, sampled at 4 times the symbolrate.

$h = \begin{bmatrix}h_{o} \\h_{1} \\\vdots \\h_{62} \\h_{63}\end{bmatrix}$Let A be a matrix defined as

$A = \begin{bmatrix}B \\B\end{bmatrix}$The received signal y is given by Y=Ah+n, where n is a vector of randomnoise values. The goal is to find a channel estimate h which minimizese ² =∥Aĥ−y∥ ²It can be shown (reference Haykin) that the optimal channel estimate isgiven byĥ=(A ^(H) A)⁻¹ A ^(H) yThe preamble sequence defined above was designed to have the importantpropertyA^(H)A=64I₆₄where I_(N) represents an N by N identity matrix. Hence,

$\begin{matrix}{\hat{h} = {\frac{1}{64}A^{H}y}} \\{= {\frac{1}{64}\begin{bmatrix}B^{H} & B^{H}\end{bmatrix}}} \\{= {{\frac{1}{64}\begin{bmatrix}B^{H} & B^{H}\end{bmatrix}}\;\begin{bmatrix}y_{1} \\y_{2}\end{bmatrix}}} \\{= {\frac{1}{64}{B^{H}\left( {y_{1} + y_{2}} \right)}}}\end{matrix}$

$\begin{bmatrix}h_{0} \\h_{1} \\h_{2} \\h_{3} \\h_{4} \\h_{5} \\h_{6} \\h_{7} \\\vdots \\h_{60} \\h_{61} \\h_{62} \\h_{63}\end{bmatrix} = {{\frac{1}{32}\begin{bmatrix}b_{0}^{*} & 0 & 0 & 0 & b_{1}^{*} & 0 & 0 & 0 & \cdots & b_{15}^{*} & 0 & 0 & 0 \\0 & b_{0}^{*} & 0 & 0 & 0 & b_{1}^{*} & 0 & 0 & \cdots & 0 & b_{15}^{*} & 0 & 0 \\0 & 0 & b_{0}^{*} & 0 & 0 & 0 & b_{1}^{*} & 0 & \cdots & 0 & 0 & b_{15}^{*} & 0 \\0 & 0 & 0 & b_{0}^{*} & 0 & 0 & 0 & b_{1}^{*} & \cdots & 0 & 0 & 0 & b_{15}^{*} \\b_{15}^{*} & 0 & 0 & 0 & b_{0}^{*} & 0 & 0 & 0 & \cdots & b_{14}^{*} & 0 & 0 & 0 \\0 & b_{15}^{*} & 0 & 0 & 0 & b_{0}^{*} & 0 & 0 & \cdots & 0 & b_{14}^{*} & 0 & 0 \\0 & 0 & b_{15}^{*} & 0 & 0 & 0 & b_{0}^{*} & 0 & \cdots & 0 & 0 & b_{14}^{*} & 0 \\0 & 0 & 0 & b_{15}^{*} & 0 & 0 & 0 & b_{0}^{*} & \cdots & 0 & 0 & 0 & b_{14}^{*} \\\vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & ⋰ & \vdots & \vdots & \vdots & \vdots \\b_{1}^{*} & 0 & 0 & 0 & b_{2}^{*} & 0 & 0 & 0 & \cdots & b_{0}^{*} & 0 & 0 & 0 \\0 & b_{1}^{*} & 0 & 0 & 0 & b_{2}^{*} & 0 & 0 & \cdots & 0 & b_{0}^{*} & 0 & 0 \\0 & 0 & b_{1}^{*} & 0 & 0 & 0 & b_{2}^{*} & 0 & \cdots & 0 & 0 & b_{0}^{*} & 0 \\0 & 0 & 0 & b_{1}^{*} & 0 & 0 & 0 & b_{2}^{*} & \cdots & 0 & 0 & 0 & b_{0}^{*}\end{bmatrix}} \cdot \left\lbrack \begin{matrix}{y_{0} + y_{64}} \\{y_{1} + y_{65}} \\{y_{2} + y_{66}} \\{y_{3} + y_{67}} \\{y_{4} + y_{68}} \\{y_{5} + y_{69}} \\{y_{6} + y_{70}} \\{y_{7} + y_{71}} \\\vdots \\{y_{60} + y_{124}} \\{y_{61} + y_{125}} \\{y_{62} + y_{126}} \\{y_{63} + y_{127}}\end{matrix} \right\rbrack}$

The noise floor estimate is computed over the 2^(nd) and 3^(rd) copiesof the training preamble and, also, over the 3^(rd) and 4^(th) copies ofthe training preamble. Two noise estimates are computed because acolliding signal that is received more than 16 symbols before or afterthe start of transmission will corrupt either the second or the lastcopy of the training preamble. If, for example, only the 3^(rd) and4^(th) copies of the training preamble were used, a colliding signalreceived more than 16 symbols after the start of a transmission couldresult in an over-estimated noise floor and a possible failure to detectthe collision. The noise vector is simply the difference between thereceived sample sequence (over some part of the training preamble) minusthe estimate of what should be received:

$\mspace{20mu}\begin{matrix}{e_{1} = {\left( {2 \cdot L_{tm}} \right) \cdot \left( {y_{1} - {\hat{y}}_{1}} \right)}} \\{= {\left( {2 \cdot L_{tm}} \right) \cdot \left( {y_{1} - {A \cdot {\hat{h}}_{1}}} \right)}} \\{= {\left( {2 \cdot L_{tm}} \right) \cdot \left( {y_{1} - {A \cdot \left( {A^{H} \cdot A} \right)^{- 1} \cdot A^{H} \cdot y_{1}}} \right)}} \\{= {\left( {{\left( {2 \cdot L_{tm}} \right) \cdot I_{128}} - {A \cdot A^{H}}} \right) \cdot y_{1}}}\end{matrix}$ $\mspace{20mu}\begin{matrix}{e_{2} = {\left( {2 \cdot L_{tm}} \right) \cdot \left( {y_{2} - {\hat{y}}_{2}} \right)}} \\{= {\left( {2 \cdot L_{tm}} \right) \cdot \left( {y_{2} - {A \cdot {\hat{h}}_{2}}} \right)}} \\{= {\left( {2 \cdot L_{tm}} \right) \cdot \left( {y_{2} - {A \cdot \left( {A^{H} \cdot A} \right)^{- 1} \cdot A^{H} \cdot y_{2}}} \right)}} \\{= {\left( {{\left( {2 \cdot L_{tm}} \right) \cdot I_{128}} - {A \cdot A^{H}}} \right) \cdot y_{2}}}\end{matrix}$    where: $\mspace{20mu}{A = \begin{bmatrix}s_{0}^{*} & 0 & 0 & 0 & s_{15}^{*} & 0 & 0 & 0 & \ldots & s_{1}^{*} & 0 & 0 & 0 \\0 & s_{0}^{*} & 0 & 0 & 0 & s_{14}^{*} & 0 & 0 & \ldots & 0 & s_{1}^{*} & 0 & 0 \\0 & 0 & s_{0}^{*} & 0 & 0 & 0 & s_{14}^{*} & 0 & \ldots & 0 & 0 & s_{1}^{*} & 0 \\0 & 0 & 0 & s_{0}^{*} & 0 & 0 & 0 & s_{14}^{*} & \ldots & 0 & 0 & 0 & s_{1}^{*} \\s_{1}^{*} & 0 & 0 & 0 & s_{0}^{*} & 0 & 0 & 0 & \ldots & s_{2}^{*} & 0 & 0 & 0 \\0 & s_{1}^{*} & 0 & 0 & 0 & s_{0}^{*} & 0 & 0 & \ldots & 0 & s_{2}^{*} & 0 & 0 \\0 & 0 & s_{1}^{*} & 0 & 0 & 0 & s_{0}^{*} & 0 & \ldots & 0 & 0 & s_{2}^{*} & 0 \\0 & 0 & 0 & s_{1}^{*} & 0 & 0 & 0 & s_{0}^{*} & \ldots & 0 & 0 & 0 & s_{2}^{*} \\\vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & ⋰ & \vdots & \vdots & \vdots & \vdots \\s_{15}^{*} & 0 & 0 & 0 & s_{14}^{*} & 0 & 0 & 0 & \ldots & s_{0}^{*} & 0 & 0 & 0 \\0 & s_{15}^{*} & 0 & 0 & 0 & s_{14}^{*} & 0 & 0 & \ldots & 0 & s_{0}^{*} & 0 & 0 \\0 & 0 & s_{15}^{*} & 0 & 0 & 0 & s_{14}^{*} & 0 & \ldots & 0 & 0 & s_{0}^{*} & 0 \\0 & 0 & 0 & s_{15}^{*} & 0 & 0 & 0 & s_{14}^{*} & \ldots & 0 & 0 & 0 & s_{0}^{*}\end{bmatrix}}$    and                                     $\mspace{20mu}{A^{H} = \begin{bmatrix}s_{0} & 0 & 0 & 0 & s_{1} & 0 & 0 & 0 & \ldots & s_{15} & 0 & 0 & 0 \\0 & s_{0} & 0 & 0 & 0 & s_{1} & 0 & 0 & \ldots & 0 & s_{15} & 0 & 0 \\0 & 0 & s_{0} & 0 & 0 & 0 & s_{1} & 0 & \ldots & 0 & 0 & s_{15} & 0 \\0 & 0 & 0 & s_{0} & 0 & 0 & 0 & s_{1} & \ldots & 0 & 0 & 0 & s_{15} \\s_{15} & 0 & 0 & 0 & s_{0} & 0 & 0 & 0 & \ldots & s_{14} & 0 & 0 & 0 \\0 & s_{15} & 0 & 0 & 0 & s_{0} & 0 & 0 & \ldots & 0 & s_{14} & 0 & 0 \\0 & 0 & s_{15} & 0 & 0 & 0 & s_{0} & 0 & \ldots & 0 & 0 & s_{14} & 0 \\0 & 0 & 0 & s_{15} & 0 & 0 & 0 & s_{0} & \ldots & 0 & 0 & 0 & s_{14} \\\vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & ⋰ & \vdots & \vdots & \vdots & \vdots \\s_{1} & 0 & 0 & 0 & s_{2} & 0 & 0 & 0 & \ldots & s_{0} & 0 & 0 & 0 \\0 & s_{1} & 0 & 0 & 0 & s_{2} & 0 & 0 & \ldots & 0 & s_{0} & 0 & 0 \\0 & 0 & s_{1} & 0 & 0 & 0 & s_{2} & 0 & \ldots & 0 & 0 & s_{0} & 0 \\0 & 0 & 0 & s_{1} & 0 & 0 & 0 & s_{2} & \ldots & 0 & 0 & 0 & s_{0}\end{bmatrix}}$A is a 128×64 matrix; and A^(H) is the Hermitian (complex-conjugate)transpose of the A matrix. S₀ . . . S₁₅ are the symbols of the trainingwhite training preamble subsequence (in order), and “*” denotes complexconjugation. Note that (A^(H). A)^(−I) is the 64×64 identity matrix(I₆₄) multiplied by the constant 1/(2*L_(trn)). In the invention,L_(trn) is 16 (symbol intervals). Writing out Y₁ and Y₂ explicitly interms of the received sample vector y (k_(offset) is the relative sampleindex from the start of transmission):

$e_{1} = {\left\lbrack {{\left( {2 \cdot L_{trn}} \right) \cdot I_{128}} - {A \cdot A^{H}}} \right\rbrack \cdot \begin{bmatrix}{y\left\lbrack {k_{offset} + {N*L_{trn}}} \right\rbrack} \\{y\left\lbrack {k_{offset} + {N*L_{trn}} + 1} \right\rbrack} \\\vdots \\{y\left\lbrack {k_{offset} + {3*N*L_{trn}} - 1} \right\rbrack}\end{bmatrix}}$$e_{2} = {\left\lbrack {{\left( {2 \cdot L_{trn}} \right) \cdot I_{128}} - {A \cdot A^{H}}} \right\rbrack \cdot \begin{bmatrix}{y\left\lbrack {k_{offset} + {2*N*L_{trn}}} \right\rbrack} \\{y\left\lbrack {k_{offset} + {2*N*L_{trn}} + 1} \right\rbrack} \\\vdots \\{y\left\lbrack {k_{offset} + {4*N*L_{trn}} - 1} \right\rbrack}\end{bmatrix}}$It turns out the matrix ((2−L_(trn))−I₁₂₈−A−A^(H)) is simply thetri-diagonal matrix:

$\left( {2 \cdot L_{trn}} \right) \cdot \begin{bmatrix}1 & 0 & 0 & 0 & \cdots & 0 & {- 1} & 0 & \cdots & 0 & 0 & 0 & 0 \\0 & 1 & 0 & 0 & \cdots & 0 & 0 & {- 1} & \cdots & 0 & 0 & 0 & 0 \\0 & 0 & 1 & 0 & \cdots & 0 & 0 & 0 & ⋰ & \vdots & 0 & 0 & 0 \\0 & 0 & 0 & 1 & \cdots & 0 & 0 & 0 & \cdots & {- 1} & \vdots & 0 & 0 \\0 & 0 & 0 & 0 & ⋰ & \vdots & 0 & 0 & \cdots & 0 & {- 1} & \vdots & 0 \\0 & 0 & 0 & 0 & \cdots & 1 & \vdots & 0 & \cdots & 0 & 0 & {- 1} & \vdots \\\vdots & 0 & 0 & 0 & \cdots & 0 & 1 & \vdots & \cdots & 0 & 0 & 0 & {- 1} \\{- 1} & \vdots & 0 & 0 & \cdots & 0 & 0 & 1 & \cdots & 0 & 0 & 0 & 0 \\0 & {- 1} & \vdots & 0 & \cdots & 0 & 0 & 0 & ⋰ & \vdots & 0 & 0 & 0 \\0 & 0 & {- 1} & \vdots & \cdots & 0 & 0 & 0 & \cdots & 1 & \vdots & 0 & 0 \\0 & 0 & 0 & {- 1} & \cdots & 0 & 0 & 0 & \cdots & 0 & 1 & \vdots & 0 \\0 & 0 & 0 & 0 & ⋰ & \vdots & \vdots & \vdots & \cdots & 0 & 0 & 1 & \vdots \\0 & 0 & 0 & 0 & \cdots & {- 1} & 0 & 0 & \cdots & 0 & 0 & 0 & 1\end{bmatrix}$So, the noise estimates can be reduced to the following simplecalculations:e ₁(k)=y(k+k _(offset) +N·L _(trn))−y(k+k _(offset)+2·N·L _(trn)), k=0 .. . N·L _(trn)−1e ₂(k ₎ =y(k+k _(offset)+2·N·L _(trn))−y(k+k _(offset)+3·N·L _(trn)),k=0 . . . N·L _(trn)−1In a proposed embodiment, the error vectors are S0.9 values. If overflowof any intermediate computation occurs, a collision is declared.

In accordance with the present invention, a method and apparatus isprovided for the computation of the variance of each noise estimate.

$\eta_{1} = {\frac{1}{\left( {N \cdot 2 \cdot L_{trn}} \right)} \cdot {\sum\limits_{k = 0}^{N \cdot L_{trn}}{{mag\_ approx}\left( {e_{1}(k)} \right)^{2}}}}$$\eta_{2} = {\frac{1}{\left( {N \cdot 2 \cdot L_{trn}} \right)} \cdot {\sum\limits_{k = 0}^{N \cdot L_{trn}}{{mag\_ approx}\left( {e_{2}(k)} \right)^{2}}}}$For the fixed-point calculations in one proposed embodiment, rounding isused in the right shift (of 7 bits); the output is a 19-bit quantity.However, any datapath width may be used. If η₁<η₂, the η=η₁ and thechannel estimate is computed using the 2^(nd) and 3^(rd) copies of thetraining preamble; otherwise η=η₂ and the channel estimate is computedusing the 3^(rd) and 4^(th) copies of the training preamble. Theresulting value η is then clipped to be within η_(low) and η_(high) (thelow and high noise points). In a proposed embodiment, both of thesevalues are 9-bit “dB” values that are used to control the range ofallowable noise variance estimates, i.e.,:η>η_(high)→η=_(high)η<η_(low)→η=η_(low)If |10*log₁₀(η₁)−10*log₁₀(η₂)|>cd_threshold_1, then a collision isdeclared. This test will very rarely pass when a collision has notoccurred and catches the case of a colliding signal received more than16 symbols before or after the start of transmission.

The last aspect of the collision detection is a unique field match. Inaccordance with the present invention, a transmitter sends bursts whichhave a unique source (“SRC”) address. In addition, there are otherheader fields which may be useful in a unique symbol template. In thefollowing sections, a preferred embodiment, using the “SRC” field as theunique symbol template, is described. However, combinations of otherfields, for example, SRC and destination address (“DST”), the “scramblerinitialization” (SI) field and the SRC and DST fields, and the SI andSRC fields, may be used. In accordance with the present invention, theSRC field is included in the template. Referring back to FIGS. 6 and 8,the transmitted burst (frame) format is illustrated. “DA” and “SA”correspond to “DST” and “SRC” in the remaining text. The SRC sampletemplate (as distinguished from the SRC symbol template) is a sequenceof sample values at four times the symbol rate (T/4) which spans the 24symbols (12 bytes) of the source address and 4 additional symbols of“guard” following the SRC field. The “guard” portion accounts for theprecursor of the channel impulse response. If only the SRC field isconsidered unique (as in the preferred embodiment), the SRC sampletemplate is computed as the linear convolution of the gth symbol of theDST field through the four symbols following the SRC field with thechannel estimate. Other sample templates are possible, including anywhich are generated as the linear convolution of a unique symboltemplate and the channel estimate. The channel estimate is computedeither from the 2^(nd) and 3^(rd) or from the 3^(rd) and 4^(th) copiesof the preamble, as described above. In the preferred embodiment, thelinear convolution is computed in three pieces, and the three componentsof the template are summed to produce the result. The first component,the destination on symbol component, is computed as follows (63×63matrix by 63×1 column vector):

$y_{DST} = {\begin{bmatrix}h_{63} & h_{62} & h_{61} & h_{60} & h_{59} & h_{58} & h_{57} & h_{56} & \cdots & h_{3} & h_{2} & h_{1} & h_{0} \\0 & h_{63} & h_{62} & h_{61} & h_{60} & h_{59} & h_{58} & h_{57} & \cdots & h_{4} & h_{3} & h_{2} & h_{1} \\0 & 0 & h_{63} & h_{62} & h_{61} & h_{60} & h_{59} & h_{58} & \cdots & h_{5} & h_{4} & h_{3} & h_{2} \\0 & 0 & 0 & h_{63} & h_{62} & h_{61} & h_{60} & h_{59} & \cdots & h_{6} & h_{5} & h_{4} & h_{3} \\0 & 0 & 0 & 0 & h_{63} & h_{62} & h_{61} & h_{60} & \cdots & h_{7} & h_{6} & h_{5} & h_{4} \\0 & 0 & 0 & 0 & 0 & h_{63} & h_{62} & h_{61} & \cdots & h_{8} & h_{7} & h_{6} & h_{5} \\0 & 0 & 0 & 0 & 0 & 0 & h_{63} & h_{62} & \cdots & h_{9} & h_{8} & h_{7} & h_{6} \\0 & 0 & 0 & 0 & 0 & 0 & 0 & h_{63} & \cdots & h_{10} & h_{9} & h_{8} & h_{7} \\\vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & \vdots & ⋰ & \vdots & \vdots & \vdots & \vdots \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & \cdots & h_{63} & h_{62} & h_{61} & h_{60} \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & \cdots & 0 & h_{63} & h_{62} & h_{61} \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & \cdots & 0 & 0 & h_{63} & h_{62} \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & \cdots & 0 & 0 & 0 & h_{63}\end{bmatrix} \cdot \left\lbrack \begin{matrix}0 \\0 \\0 \\{s_{dst}(9)} \\0 \\0 \\0 \\\vdots \\{s_{dst}(23)} \\0 \\0 \\0\end{matrix} \right\rbrack}$The “guard” symbol component is computed as follows (16×16 matrix by16×1 column vector):

$y_{GRD} = {\begin{bmatrix}h_{0} & 0 & 0 & 0 & \cdots & 0 \\h_{1} & h_{0} & 0 & 0 & \cdots & 0 \\h_{2} & h_{1} & h_{0} & 0 & \cdots & 0 \\h_{3} & h_{2} & h_{1} & h_{0} & \cdots & 0 \\\vdots & \vdots & \vdots & \vdots & ⋰ & \vdots \\h_{15} & h_{14} & h_{13} & h_{12} & \cdots & h_{0}\end{bmatrix} \cdot \begin{bmatrix}{s_{rsv}(0)} \\0 \\0 \\0 \\\vdots \\{s_{rsv}(3)} \\0 \\0 \\0\end{bmatrix}}$The source symbol component is computed as follows (112×96 matrix by96×1 column vector):

$y_{SRC} = {\begin{bmatrix}h_{0} & 0 & 0 & 0 & 0 & 0 & 0 & \cdots & 0 & 0 & 0 & 0 \\h_{1} & h_{0} & 0 & 0 & 0 & 0 & 0 & \cdots & 0 & 0 & 0 & 0 \\h_{2} & h_{1} & h_{0} & 0 & 0 & 0 & 0 & \cdots & 0 & 0 & 0 & 0 \\h_{3} & h_{2} & h_{1} & h_{0} & 0 & 0 & 0 & \cdots & 0 & 0 & 0 & 0 \\h_{4} & h_{3} & h_{2} & h_{1} & h_{0} & 0 & 0 & \cdots & 0 & 0 & 0 & 0 \\h_{5} & h_{4} & h_{3} & h_{2} & h_{1} & h_{0} & \cdots & \cdots & 0 & 0 & 0 & 0 \\\vdots & \vdots & \vdots & \vdots & \vdots & \vdots & ⋰ & \vdots & \vdots & \vdots & \vdots & \vdots \\h_{63} & h_{62} & h_{61} & h_{60} & h_{59} & h_{58} & \cdots & h_{0} & 0 & 0 & 0 & 0 \\0 & h_{63} & h_{62} & h_{61} & h_{60} & h_{59} & \cdots & h_{1} & h_{0} & 0 & 0 & 0 \\0 & 0 & h_{63} & h_{62} & h_{61} & h_{60} & \cdots & h_{2} & h_{1} & h_{0} & 0 & 0 \\\vdots & \vdots & \vdots & ⋰ & ⋰ & ⋰ & ⋰ & ⋰ & ⋰ & ⋰ & ⋰ & 0 \\0 & 0 & 0 & \cdots & h_{63} & h_{62} & h_{61} & h_{60} & \cdots & h_{2} & h_{1} & h_{0} \\0 & 0 & 0 & \cdots & 0 & h_{63} & h_{62} & h_{61} & \cdots & h_{3} & h_{2} & h_{1} \\0 & 0 & 0 & \cdots & \vdots & \vdots & ⋰ & \vdots & \vdots & \vdots & \vdots & \vdots \\0 & 0 & 0 & \cdots & 0 & 0 & \cdots & h_{63} & \cdots & h_{18} & h_{17} & h_{16}\end{bmatrix} \cdot \left\lbrack \begin{matrix}{s_{src}(0)} \\0 \\0 \\0 \\{s_{src}(1)} \\0 \\0 \\0 \\\vdots \\{s_{src}(23)} \\0 \\0 \\0\end{matrix} \right\rbrack}$The template signal is computed as:

${\hat{y}(k)} = \left\{ \begin{matrix}{{{y_{SRC}(k)} + {y_{DST}(k)}},{k = {{0\mspace{20mu}\ldots\mspace{14mu}{N \cdot L_{trn}}} - 2}}} \\{{y_{SRC}(k)},{k = {{N \cdot L_{trn}} - {1\mspace{14mu}\ldots\mspace{14mu}{N \cdot L_{src}}} - 1}}} \\{{{y_{SRC}(k)} + {y_{GRD}\left( {k - {N \cdot L_{trn}}} \right)}},{k = {{{N \cdot L_{src}}\mspace{14mu}\ldots\mspace{14mu}{N \cdot \left( {L_{src} + L_{grd}} \right)}} - 1}}}\end{matrix} \right.$Graphically, the template is constructed as set forth in FIG. 64. TheSRC match error sequence is then just the 112-point sequence defined by:e(k)=ŷ(k)−(2·L _(trn))·y(k+k _(offset) +N·(4· _(—) L _(trn) +L _(dst))),k=0 . . . N·(L _(src) +L _(grd))−1The template is built in the following order:

-   1. initialize the template to the negation of the input sample    sequence (shifted left by 6 bits)-   2. add the y_(DST) term-   3. add the y_(GRD) term-   4. add the y_(SRC) term    If overflow of the template accumulator occurs at any point in these    calculations, then a collision is declared. The size of the    accumulator may vary in this invention, but the preferred embodiment    uses r+5 bits, where r is the number of analog-to-digital converter    output bits. The signed SRC match error is the input to a magnitude    approximation, computed as max(x_(j), x_(q))+½* min (x_(j), x_(q)),    where x_(i) is the in-phase component of the complex sample and    x_(q) is the quadrature component, with rounding. Referring to FIG.    65 the maximum over the magnitude of all N*(L_(trn)+L_(grd))=112    error output samples is found; this is called |e_(max)|. The maximum    error between the received SRC field samples and the estimated SRC    field samples is compared against a threshold which varies with the    noise floor estimate. The datapath widths are those used in the    preferred embodiment, but other datapath widths can be implemented    in accordance with the present invention. If a “no match” result is    returned, a collision occurred. In one embodiment, k_(offset) is an    8-bit signed value (in units of T/4 samples) that indicates where    the hybrid leakage is received relative to the start of    transmission. k_(offset)+76 samples would be the index on which the    “start” signal would be asserted if the carrier sense were running.    It must be writable from the external interface. In one embodiment,    the 20*log 10(.) function is implemented with two 16-entry look-up    tables, a coarse 20*log 10(.) table and a fine 20*log 10(.) table.    Referring to FIGS. 66 a and 66 b, the table set forth in FIG. 66 a    contains unsigned 7.2 values and the table set forth in FIG. 66 b    contains unsigned 3.2 values. The algorithm is:-   1. ×=max (x, 2)-   2. Find the most significant non-zero bit in the input, x. Call the    position of this bit (0 . . . 15) b.-   3. d₁=coarse_tbl[b]-   4. if b>3, k=(x−2^(b))<<(b−4), else k=(x−2^(b))>>(4−b).-   5. d₂=fine_tbl[k]-   6. output=d₁+d₂    The output is in 7.2 format, representing up to 96 dB of dynamic    range with up to 0.25 dB resolution.

Gain Estimation Circuit for Burst Modem

Now turning to the gain estimation circuit aspect of the presentinvention, in digital burst modem receiver designs, either ananalog-to-digital converter (ADC) sufficient to meet the dynamic rangerequirements of the signal propagation path is used, or alower-precision ADC may often be used in conjunction with avariable-gain amplifier. For many applications, e.g. phoneline,powerline, or wireless networking, the dynamic range requirements arelarge (often greater than 60 dB), because the path propagation loss canvary between 0 dB and 40 to 70 dB. To meet system requirements with theformer approach, a very large and expensive ADC would be required.Therefore, designers often choose a lower-precision ADC with avariable-gain amplifier and some control circuit. In accordance with thepresent invention, a circuit is provided which estimates the requiredgain from a carefully-designed preamble at the start of a burst(packet). A preamble of M*N symbol duration is presupposed. The preambleis structured so that M identical copies of an N-symbol subsequence aretransmitted sequentially. Each N-symbol subsequence has the property ofbeing spectrally white (i.e. nonzero cyclic autocorrelation at only oneof the N possible lag values). In accordance with the invention, thestart of a burst (packet) is detected at the receiver. This function isusually referred to as “carrier sense”. The circuit set forth hereintakes a binary start-of-frame/no-start-of-frame indication from thecarrier sense. The circuit in accordance with the present invention hastwo subcircuits. The first estimates the received signal power over anN-symbol window. This subcircuit runs continuously. The second estimatesthe required gain to maximize signal-to-noise ratio, given the powerestimate, and the binary indication from carrier sense. It runs onlyafter carrier sense has indicated the start of burst, and it ceasesoperation after it computes a gain code, until the next start of burstindication. The input to the power estimation circuit is a k-bitreceived signal magnitude value. This value is the output the ADC withlittle or no frequency-selective (e.g. low-pass or band-pass) filtering.The power accumulator, which is 2*k+log 2(L*N) unsigned bits, isinitialized to zero. N is the number of symbols in one copy of thepreamble, and L is the ADC oversampling factor (sampling frequencydivided by symbol frequency). The oversampling factor L is chosen to besufficiently-large to avoid aliasing. For the first L*N samples afterinitialization, the square of each new k-bit magnitude value (_(nk)) isadded to the accumulator. No values are subtracted from the poweraccumulator. On and following the (N*L−1)^(th) sample afterinitialization, the power accumulator calculation becomes:pwr_acc=pwr_acc+Z _(k) ² −Z _(k−nk−(N*L)) ².A block diagram of a possible embodiment, in which k=7 bits, is shown inFIG. 67. The gain code update is triggered by the start of burstindication from carrier sense. At this point, the upper 2*k bits of thepower accumulator (rounded up) are passed to the gain code calculation.The combination of the preamble structure and the power estimationcircuit yields a nearly-constant power estimate for anyless-than-N-symbol shift of the start of burst indication. This propertyallows the start of burst indication to be somewhat inaccurate (becauseof channel characteristics, etc.). The gain control block operates onceon every start of burst transition and takes the power estimator outputsample (2*k bits) as its input. One commercially-available embodimentprovides a control range of 22.5 dB for the variable second-stageamplifier with a step size of roughly 0.75 dB, but many other controlranges and step sizes are possible. There are two programmable tablesand a programmable back_off value. The programmable back-off valueallows fixed gain variations due to particular implementations and alsoallows variations in the peak-to-average ratio of the received signal tobe tolerated by this circuit without unacceptable saturation of thereceiver. The first programmable table, rough_gain_table, is a r0-entryby 10-bit array of codes mapping to very coarse gain values. In oneembodiment, it is an 8-entry table with 8-bit values corresponding togain values of {0, 3, 6, 9, 12, 15, 18, 21, 24} dB. The second table,fine_gain_table, is r1-entry by I1-bit array of codes mapping to arefinement of those coarse values. One embodiment might use a 4-entry by8-bit array of codes mapping to gain values 0.75 dB, 1.5 dB, 2.25, and 3dB. The 10-bit back-off value allows for fixed gain variations duringsystem bring-up and peak-to-average ratio variations. There is also ar2-entry by 12-bit sparse fixed table, called the fine_log_map, whichmaps integers 0 . . . (r2−1) to the corresponding fine_gain_table bin.In one embodiment, this may be a 32-entry by 4-bit sparse fixed table,which maps integers 0 . . . 31 to a corresponding fine_gain_table bin.It could contain the values {0, 0, 0, 0, 0, 0, 0, 1, 1, 1, 1, 1, 1, 1,2, 2, 2, 2, 2, 2, 2, 2, 3, 3, 3, 3, 3, 3, 3, 3, 3, 3} in ascendingorder, as an example. The gain control calculation can be summarized inthe following steps:Set b to the minimum of 2^((k−)1)−1 and the 2*k-bitaveraged power input value; Determine the highest nonzero bit positionin b; i.e. r=max[floor(log 2(b)), log 2(r2)]; If r>log 2(r2)−1, in thelist according to additional selection criteria. The protocol toleratesloss of control frames, and provides a mechanism for retransmission oflost control frames without excessively loading the network. Theprotocol also provides a mechanism for adaptive selection of theencoding to use for transmission to a group of nodes (a “multicastgroup”). In particular, a node receiving data frames first gathersstatistics from frames sent at any encoding, and extrapolates thesestatistics to estimate the expected frame error rate of all possibleencodings. Using the estimated frame error rates, the data receivingnode computes a performance metric for all possible encodings. Using theperformance metrics, the data receiving node selects an encoding for useon the channel between the sender and the receiver to maximize networkthroughput subject to frame error rate constraints.

With regard to the Rate Negotiation and Selection Algorithms, considerthe packet data network, as described hereinabove, where a series ofpackets P₁, each composed of a fixed length header and a variable lengthpayload, are transmitted from a data sender station, A, to a datareceiver station, B. The headers are transmitted using FDQAM modulationat symbol rate S_(min), and constellation size b_(i) bits per symbol.The payloads are transmitted using QAM or FDQAM modulation at symbolrate S_(i), and constellation size b_(i) from the sets S, and R,respectively, to maximize network throughput subject to additionalconstraints, in the presence of time-varying impairments. We assume thatthere is a mechanism for error detection, but not for error correction.We also assume that there is a mechanism for B to notify A of packeterrors, and for A to retransmit such errored packets upon receivingnotification from B. The Rate Negotiation and Selection algorithms mustoperate in dynamic environments, with time varying impairments. Severalof these impairments are defined to develop the algorithms in accordancewith the present invention:

-   -   Channel Frequency Selectivity: Spectral nulls or severe        attenuation in isolated portions of the band of interest lead to        packet error rate (PER) performance that varies with symbol        rate. While we assume that the channel is quasi-static, (fixed        for the duration of several transmitted packets), it may change        between packets, P_(i).    -   Time Invariant White Noise: Noise at the input to B which has        flat Power Spectral Density (PSD) over the band of interest.        This impairment is also quasi-static.    -   Time Varying White Noise: Noise at the input to B, which has        flat PSD over the band of interest, but with level changing        between packets, P_(i), on a time scale similar to the packet        duration.    -   Periodic Impulse Noise: Periodic high amplitude impulses at the        input to B.    -   Colored Noise: Noise at the input to B, which has        frequency-dependent PSD over the band of interest.    -   This impairment may be either quasi-static, or vary on a time        scale similar to the packet duration.        The process of selecting (s_(i), b_(i)) is separated into        functions performed by the data sender, A, and the data        receiver B. B collects statistics on packets received from A,        performing a Rate Selection Algorithm, to choose (s_(i,desired);        b_(i,desired)). B transmits Rate Request Control Frames (RRCFs)        to A, typically using (s_(min), b_(min)), to signal changes in        (s_(i,desired,) b_(i,desired)). A honors B's requests, changing        (s_(i), b_(i)) to (s_(i,desired), b_(i,desired)) in response to        RRCFs from B. RRCFs are full, legal frames which are treated the        same as data frames by the MAC layer. They may be sent at a        different priority than best-effort data frames to provide        faster adaptation. RRCFs are sent when node B selects a desired        rate bin_index=2*k−1−r,b′=bitand(floor(b>>(r−log 2(r2))), 31);        else bin_index=2*k−1−log₂(r2), b′=1; bin₁₃ gain=rough_gain_table        [bin index, step . . . gain=fine_gain_table[fine_log_map [b′]];        G=max (bin_gain−step_gain−back_off, 0). The G output is        guaranteed to be an r0-bit quantity. When the carrier sense does        not indicate a start-of-burst condition (no signal present at        the receiver), G is always set to the nominal gain setting (0        dB). The computed gain code value is used to set the analog        front-end variable gain value.

As seen in FIG. 67, a power estimate is provided. Delay line 4010 is 64samples deep. 10 bits in phase and quadrature components 4012 go intodelay buffer 4010 and into multiplexer 4014. Magnitude approximator 4106provides a 10 bit output which is provided to bit selector 4018 whichtakes out 7 bits with rounding. Squaring operation 4020 is thenperformed to provide a power estimate as output. The power estimateoutput is provided to demultiplexer 4022. Accumulator 4024 takes theoutput from the demultiplexer 4024 and provides it to bit selector 4026which in turn provides an averaged power estimate 4028.

Rate Negotiation and Rate Selection

As described above, in accordance with the present invention thedynamically selecting of the encoding of data frames on a network wherenodes can transmit frames with various encodings is provided. Theencodings may vary several parameters including but not limited to thenumbers of bits per symbol, the number of symbols per second, or thefrequency band(s) used. A node receiving data frames makes adetermination about which encodings are appropriate for use on thechannel between the sender and the receiver. Multiple encodings may beselected. The data frame receiver then notifies the data frame sender ofthe encoding selections, with an indication of the relative usability ofthe selected encodings, via a control frame. The sender is free to useany of the specified encodings, or may use one not included whichdiffers from the current rate, with some backoff algorithm to preventflooding the network with rate change requests and to allow time forrate info to arrive and get processed at A. Various backoff algorithmsmay be employed including: (1) truncated binary exponential backoff(BEB) with backoff based on received frame count, (2) truncated BEB withbackoff based on time weighted truncated backoff (based on receivedframe count or time) where the backoff function is dependent on thedesired and current payload encoding rates, and need not be a binaryexponential function.

Rate Selection refers to the algorithm by which B chooses(S_(i,desired), b_(i,desired)). Each of the algorithms presented usesome or all of the following input statistics upon receiving packetP_(i), (squared error refers to squared error refers to squared decisionpoint error): Header rate, (s_(min),b_(min)); Header error indicator,X_(hdr,i) ε{0,1}, O indicates error-free header, 1 indicates headererror; Header sum of squared error, ε_(hdr,i); Header maximum squarederror, E_(hdr,1); Header length symbols), n_(hdr); Payload rates,(s_(i), b_(i)) Payload error indicator, X_(hdr,i) ε{0,1}, 0 indicateserror-free payload, 1 indicates payload error; Payload sum of squarederror, ε_(pld,i); Payload maximum squared error, E_(pdi,i); Payloadlength (symbols), n_(pld,i); FSE power for each symbol rate in S,P_(FSE,s,i;) and Normalized, per-symbol ISI power estimate for eachsymbol rate in S, P_(ISI,s,i). Given these input statistics, eachalgorithm maintains state variables, performing computations based onthe input statistics and state variables, first to select the newdesired constellation size from R_(s) for each symbol rate in S, then toselect the new desired symbol rate from all those in S. Two algorithmsare presented, requiting different amounts of state storage andcomputation: (1) Mean Squared Error Algorithm and (2) Maximum SquaredError Algorithm. For the purpose of constellation size selection, weinitially assume that only a single symbol rate, s, is underconsideration, and that s_(i)=x for all i.

With regard to the Mean Squared Error Algorithm, error rates ofcandidate constellations are estimated, selecting constellation tomaximize throughput subject to maximum length packet, maximum PERconstraint. If we assume that: probability of symbol error isindependent from symbol to symbol, hence:

${{PER}\left( {{SNR},b} \right)} \equiv {1 - \left( {1 - {{SER}\left( {{SNR},b} \right)}} \right)^{\frac{N_{\max}}{b}}}$where: $\begin{matrix}{N_{\max} = {{maximum}\mspace{14mu}{packet}\mspace{14mu}{length}\mspace{14mu}({bits})}} \\{b = \text{candidate constellation size (bits per symbol)}} \\{{SNR} = {\text{symbol decision point signal to noise ratio, normalizedby loss in mean symbol energy of constellation size}\text{b}\text{relative to constellation size}b_{\min}}} \\{{SER} = {{symbol}\mspace{14mu}{error}\mspace{14mu}{rate}}}\end{matrix}$Noise is additive, white, and Gaussian, hence:

${{SER}\left( {{SNR},b} \right)} \equiv \left\{ {{\begin{matrix}{2 \cdot {Q\left( \sqrt{2 \cdot \frac{SNR}{b}} \right)} \cdot \left( {1 - {\frac{1}{2}{Q\left( \sqrt{2 \cdot \frac{SNR}{b}} \right)}}} \right)} & {{b = 2},{QAM}} \\{2 \cdot {Q\left( {\sqrt{2 \cdot {SNR}} \cdot {\sin\left( \frac{\pi}{2^{b}} \right)}} \right)}} & {{b = 3},{PSK}} \\{1 - \left( {1 - {2 \cdot {Q\left( \sqrt{3 \cdot \frac{1}{2^{b - 1}} \cdot {SNR}} \right)}}} \right)^{2}} & {{b > 3},{odd},{QAM}} \\{1 - \left( {1 - {2 \cdot \left( {1 - \frac{1}{\sqrt{2^{b}}}} \right) \cdot {Q\left( \sqrt{3 \cdot \frac{1}{2^{b - 1}} \cdot {SNR}} \right)}}} \right)^{2}} & {{b > 3},{even},{QAM}}\end{matrix}{where}\text{:}{Q(x)}} = {\frac{1}{2\pi}{\int_{x}^{\infty}{{\mathbb{e}}^{- \frac{x^{2}}{2}}{\mathbb{d}x}}}}} \right.$(see J. G. Proakis, Communications Systems Engineering, section 9.2–9.3,1994). Then we can precompute SNR thresholds, SNR_(min,b) for eachconstellation size b in R, such that: PER(SNR_(min,b))=PER_(max). Hence,PER<PER_(max) for SNR>SNR_(min.b) is monotonically increasing with b.However, in the event that either the independent SER assumption or thewhite noise assumption are invalid, SNR_(min,b) is computed by differentmeans, such that PER<PER_(max) is still satisfied for SNR>SNR_(min,b.)Following each received packet P_(i), with error-free header,X_(hdr,i)=0, and error free payload, X_(pld,i)=0, an SNR estimate,{overscore (SNR)}_(i) is computed. A sliding-window average of length Nis used to bound maximum response time to N received packets, so stateis needed to store the last N−1 values of ε_(hdr), ε_(pld), and n_(pld:)

${\overset{\_}{SNR}}_{i} \equiv {E_{s,b_{\min}} \cdot \left( \frac{{N \cdot n_{hdr}} + {\sum\limits_{j = {i - {({N - 1})}}}^{i}n_{{pld},j}}}{\sum\limits_{j = {i - {({N - 1})}}}^{i}\left( {ɛ_{{hdr},j} + ɛ_{{pld},j}} \right)} \right)}$

where E_(s,b) _(min) is the mean energy per symbol for constellationsize b_(min).

If the implementation is memory constrained, a single-pole average isused and state is only needed to store the last value of the erroraverage, {overscore (ε)}_(i−1):

${\overset{\_}{ɛ}}_{i,{hdr}} \equiv {{{\overset{\_}{ɛ}}_{i - 1} \cdot \left( {1 - \frac{1}{\tau}} \right)^{n_{{hdr},i}}} + {\left( \frac{ɛ_{{hdr},i}}{n_{{hdr},i}} \right) \cdot \left( {1 - \left( {1 - \frac{1}{\tau}} \right)^{n_{{hdr},i}}} \right)}}$${\overset{\_}{ɛ}}_{i} \equiv {{{\overset{\_}{ɛ}}_{i,{hdr}} \cdot \left( {1 - \frac{1}{\tau}} \right)^{n_{{pld},i}}} + {\left( \frac{ɛ_{{pld},i}}{n_{{pld},i}} \right) \cdot \left( {1 - \left( {1 - \frac{1}{\tau}} \right)^{n_{{pld},i}}} \right)}}$${\overset{\_}{SNR}}_{i} \equiv {E_{s,b_{\min}} \cdot {\overset{\_}{ɛ}}_{i}^{- 1}}$where τ is a time constant describing the weighting of new measurementsin the average. Note that the implementation above inherently weightspayload measurements more strongly that header measurements. In caseswhere it is desirable to place stronger weight on header measurements,the following substitutions are made for n_(hdr) and n_(pld,i):n′_(hdr)>n_(hdr) and n′_(pld,i)<n_(pld,i). Note that the power andmultiply operations above can be implemented with a lower accuracy fixedprecision approach, without significant performance impact. Followingeach received packet P_(i), {overscore (SNR)}_(i) is compared to thethreshold set SNR_(min,b) to choose the new constellation size,b_(i+1,desired):

${b_{{i + 1},{up}} = {\max\limits_{b \in B}(b)}},{{{subject}\mspace{14mu}{to}\mspace{14mu}{\overset{\_}{SNR}}_{i}} > {{SNR}_{\min,b} + \Delta}}$${b_{{i + 1},{down}} = {\max\limits_{b \in B}(b)}},{{{subject}\mspace{14mu}{to}\mspace{14mu}{\overset{\_}{SNR}}_{i}} > {SNR}_{\min,b}}$b_(i + 1, desired) = max (b_(i + 1, up), min (b_(i, desired), b_(i + 1, down)))The offset, Δ, provides decision stability when {overscore (SNR)}_(i)has value near a threshold SNR_(min,b).

With regard to the Maximum Squared Error Algorithm, error rates ofcandidate constellations are estimated, selecting the constellation tomaximize throughput subject to minimum percent improvement constraint.Assume that a symbol error occurs for candidate constellations size b,if the decision point error exceeds half the constellation minimumdistance, d_(min,b). Also, assume that d_(min.b) is monotonicallydecreasing with b. Given a series of n_(i) symbols transmitted usingconstellation b_(i), with error indicator χ_(i)=1, we can declare thatat least one symbol error would have occurred as well for n_(i) symbolstransmitted using constellation b≧b_(i). Given a series of n_(i) symbolstransmitted using constellation b_(i), with maximum squared error E_(i),and error indicator χ_(i)=0, we can declare whether at least one symbolerror would have occurred instead for n_(i) symbols transmitted usingconstellation b>bi, if E_(i)>(d_(min.b)/2)². Thus, we say symbol errorsare “observable” for candidate constellation sizes b≦b_(i). Followingeach received packet P_(i), the average payload length estimate, N_(i),is updated:if χ_(hdr,i)=0. and χ_(pld,i)=0:N_(i)=N_(i−1)·(1−l/τ)+n_(pld,i) ·b _(i) ·l/τotherwiseN_(i)=N_(i−1)where τ is a time constant describing the weighting of new measurementsin the average. Following each received packet p_(i), thelength-independent success rate, Z_(1b.i), is updated for allconstellation sizes b>bmin:

if  χ_(hdr, i) = 0: $Z_{{1b},i} = \left\{ {{\begin{matrix}1 & {{{for}\mspace{14mu} b} = b_{\min}} \\{{Z_{{1b},{i - 1}} \cdot \left( {1 - \frac{1}{\tau}} \right)} + \frac{1}{\tau}} & {{{when}\mspace{14mu} E_{{hdr},i}} < \left( \frac{d_{\min,b}}{2} \right)^{2}} \\{Z_{{1b},{i - 1}} \cdot \left( {1 - \frac{1}{\tau}} \right)} & {{{when}\mspace{14mu} E_{{hdr},i}} \geq \left( \frac{d_{\min,b}}{2} \right)^{2}}\end{matrix}\text{otherwise:}Z_{{1b},i}} = Z_{{1b},{i - 1}}} \right.$Note that Z_(1b.i) cannot be updated when _(χhdr,i)=1 if we assume thatthe header contains the packet source address, since B cannot be certainthat p_(i) was transmitted by A. Thus, Z_(1b,i) is actually notobservable for b=b_(min). Next, the conditional length-dependent successrate, Z_(2b.i,) and the corresponding average measurement length,L_(b.1,) for observable constellation sizes b≧b_(i), are updated:

${{{if}\mspace{14mu}\chi_{{hdr},i}} = 0},{{{{{and}\mspace{14mu} E_{{hdr},i}} < \left( \frac{d_{\min,\; b}}{2} \right)^{2}}:Z_{{2b},i}} = \left\{ {{\begin{matrix}{{Z_{{2b},{i - 1}} \cdot \left( {1 - \frac{1}{\tau}} \right)} + \frac{1}{\tau}} & {{{when}\mspace{14mu}\chi_{{pld},i}} = {{0\mspace{14mu}{and}\mspace{14mu} E_{{pld},i}} < \left( \frac{d_{\min,\; b}}{2} \right)^{2}}} \\{Z_{{2b},{i - 1}} \cdot \left( {1 - \frac{1}{\tau}} \right)} & {{{when}\mspace{14mu}\chi_{{pld},i}} = {1\mspace{14mu}{or}}}\end{matrix}\mspace{329mu}\left( {\chi_{{pld},i} = {{0\mspace{14mu}{and}\mspace{14mu} E_{{pld},i}} \geq \left( \frac{d_{\min,\; b}}{2} \right)^{2}}} \right)L_{b,i}} = {{{L_{b,{i - 1}} \cdot \left( {1 - \frac{1}{\tau}} \right)} + {{n_{{pld},i} \cdot \frac{1}{\tau}}\text{otherwise:}Z_{{2b},i}}} = {Z_{{2b},{i - 1}}{L_{b,i} = L_{b,{i - 1}}}}}} \right.}$Next, the aggregate success rate estimate, Z_(b,i,) is computed:

$Z_{b,i} = {Z_{{1b},i} \cdot Z_{{2b},i}^{(\frac{\frac{N_{i}}{b}}{L_{b,i}})}}$Next, the expected time-on-medium metric, M_(b,i,) is computed:

$M_{b,i} = {{\frac{1}{Z_{b,i}} \cdot \left( {\rho + \frac{N_{i}}{s \cdot b}} \right)} + {\frac{1}{Z_{b,i}} \cdot \left( {\frac{1}{Z_{b,i}} - 1} \right) \cdot \rho}}$where p is the average packet overhead in seconds, including contentiontime and fixed header duration, and s is the symbol rate underconsideration. Note that 1/z_(b,i,) represents the average number oftimes that a packet must be transmitted until it is successfullyreceived, assuming each packet is received successfully with independentprobability Z_(b,i). Also, the second term in the expression for M_(b,i)accounts for the time occupied by B's retransmission requests to A,assuming a symmetric probability of success. While this assumption isnot strictly valid, the retransmission term is simply intended toprovide stronger dependence on Z_(1b.i) in the case where packets arepredominantly short, (Ni/s·b˜p). Next, the hold count h_(b,i) isupdated:

$h_{b,i} = \left\{ \begin{matrix}0 & {{{where}\mspace{14mu} M_{b,i}} \geq {\left( {1 - \Delta_{benefit}} \right) \cdot M_{b_{1},i}}} \\{h_{b,{i - 1}} + 1} & {{{where}\mspace{14mu} M_{b,i}} < {\left( {1 - \Delta_{benefit}} \right) \cdot M_{b_{1},i}}}\end{matrix} \right.$where Δ_(benefits), the minimum percent improvement in throughputrequired to justify a rate change. Finally, the new constellation size,b_(i+1.desired), is Chosen·b_(i+1.desired)=b, such that

${M_{b,i} = {\min\limits_{b\; ɛ\; B}\left( M_{b,i} \right)}},{{{subject}\mspace{14mu}{to}\mspace{14mu} h_{b,i}} \geq h_{\min}}$where h_(min) is the minimum number of consecutive packets for whichconstellation size b must provide at least Δ_(benefit) percentimprovement in throughput to justify a rate change.

It should be noted that decreasing the time constant, r, decreases theresponse time to a step change in input, but also decreases theeffective maximum, measurable, non-unity value of Z_(b,i), Z_(b.MAX):Z _(b,MAX)≡1−¹/_(τ)If it is assumed that applications perform poorly with PER greater thanPER_(max), τ is chosen large enough to measure the value Z_(b,max) ofinterest, 1−PER_(max). Thus τ˜1/PER_(max). Selecting τ by this methodfor small PER_(max) can lead to an unacceptably long response time, onthe order of hundreds of packets or more. In this case, the sensitivityto PER can be increased while maintaining a short response time, byusing an “effective” constellation minimum distance instead of theactual minimum distance described earlier:d_(min,b, effective)=d_(min,b+1). Since Z_(2b,i) is not observable forcandidate constellation sizes b<b_(i), there are some combinations ofpast input and current input that prevent the above algorithm fromselecting b<b_(i), even though higher throughput would be achieved. Thisis not the case when the current impairments are non-impulsive, sincethese impairments can still be observed via Z_(1b.i). However, if thecurrent impairments are impulsive or intermittent, they are less likelyto occur during a header, and Z_(1b.i) will be largely unaffected. Underthese conditions, b_(i.desired) is not likely to change unless enoughpayload errors occur to reduce Z_(2b.i) for the observable rates,relative to the unobservable rates. Since many applications aresensitive to bursts of consecutive errors, and a burst of errors islikely to occur before b_(i,desired) changes by the above mechanism, itis desirable to enforce a maximum tolerated number of consecutivepayload errors, after which b_(i,desired) is set to b_(probe). A naturalchoice for b_(probe) is b_(min), rendering all b in R observable. Moregenerally, b_(probe) is chosen to be b_(i.desired)−K, allowing avariable number of constellation sizes b to become observable. Thenb_(i.desired) remains set to b_(probe) until a minimum number of packetsare received using constellation size b_(probe). After this condition issatisfied, b_(i.desired) is selected as described earlier, until themaximum tolerated number of consecutive payload errors is next exceeded.While b_(i.desired) can be dropped to b_(probe) to refresh statisticsfor previously unobservable rates, this can actually result in a higherPER in the presence of impairments such as periodic impulse noise.Instead, protocol support for “probing” with control packets can be usedto refresh statistics without risking data packets. In this case, Bcould compute an alternate version of the metric _(Mb,i), substitutingunity for Z_(2b.i). This constitutes an optimistic lower bound on_(Mb,i). When the optimistic bound on _(Mb,i) is sufficiently less than_(Mb,i) itself, B can request that A send periodic non-data-beatingprobe control packets using a specific constellation size, b_(probe)Again, a natural choice for b_(probe) is b_(min). Also of note is thatthe time on medium metric may be computed after several received packetsinstead of after each received packet, to limit computation.

With regard to Symbol Rate Selection, the assumption that we areconstrained to a single symbol rate is relaxed. Assume that Bdemodulates packets from A, using the MMSE FSE/DFE structure depicted inFIG. 68, for a given symbol rate s. Assuming a fixed channel response,h_(k), fixed MMSE FSE and DFE coefficients for a given symbol rate,w_(s,k) and b_(s,k), and additive white Gaussian noise, n_(k), the meansquared error expected for symbol rate s, MSE_(s), is related to thatexpected for symbol rate s_(min), MSEs_(min), as follows:

$\frac{{MSE}_{s}}{{MSE}_{s_{\min}}} \cong \frac{{\left( {{SNR}_{s_{\min}}^{- 1} - P_{{ISI},s_{\min}}} \right) \cdot \frac{s}{s_{\min}} \cdot \frac{P_{{FSE},s}}{P_{{FSE},s_{\min}}}} + P_{{ISI},\; s}}{{SNR}_{s_{\min}}^{- 1}}$$P_{{FSE},s} = {\sum\limits_{k = 0}^{{N_{f} \cdot L} - 1}{w_{s,k}}^{2}}$$P_{{FSE},s_{\min}} = {\sum\limits_{k = 0}^{{N_{f} \cdot L} - 1}{w_{s_{\min},k}}^{2}}$$P_{{ISI},s} = {\sum\limits_{j = 0}^{N_{f} - 1}{{\sum\limits_{k}{h_{k} \cdot w_{s,{j - k}}}}}^{2}}$$P_{{ISI},s_{\min}} = {\sum\limits_{j = 0}^{N_{f} - 1}{{\sum\limits_{k}{h_{k} \cdot w_{s_{\min},{j - k}}}}}^{2}}$Furthermore, the maximum squared error over a set of n_(i) symbolsexpected for symbol rate s, E_(s,i), is related to that expected forsymbol rate s_(min), Es_(min) i as follows:

$\frac{E_{s,i}}{E_{s_{\min},i}} \cong \frac{{MSE}_{s,i}}{{MSE}_{s_{\min},i}}$Since the channel response, FSE and DFE coefficients, and noise levelmay vary from packet to packet, the squared error ratio estimator,λ_(s,i,) is introduced:

$\lambda_{s,i} \equiv {{\lambda_{s,{i - 1}} \cdot \left( {1 - \frac{1}{\tau}} \right)} + {\lambda_{new} \cdot \frac{1}{\tau}}}$where:$\lambda_{new} \equiv \frac{{\left( {{SNR}_{s_{\min},i}^{- 1} - P_{{ISI},s_{\min},i}} \right) \cdot \frac{s}{s_{\min}} \cdot \frac{P_{{FSE},s,i}}{P_{{FSE},s_{\min},i}}} + P_{{ISI},s,i}}{{SNR}_{s_{\min},i}^{- 1}}$${SNR}_{s_{\min},i} \equiv {E_{s,b_{\min}} \cdot \frac{n_{hdr}}{ɛ_{{hdr},i}}}$

-   -   and E_(s,b) _(min) is the mean energy per symbol for        constellation size b_(min).        With regard to the Mean Squared Error Algorithm, the method for        computing {overscore (SNR)}_(i) described earlier is repeated ,        but with the following substitution for ε_(pld,i):        ε′_(pld,i)≡ε_(pld,i)·λ_(s) _(i) _(,i) ⁻¹        b_(s,i+1,desired), as described earlier for b_(i+1,desired), is        chosen, but independently for each symbol rate s in S, using the        following substitution for SNR _(i):        SNR _(s,i) =SNR _(i)·λ_(s,i):        For a given symbol rate s>s_(min), if {overscore        (SNR)}_(s,i)<_(SNRmin,b) for all b in R, then        b_(s,i+1,desired)=0 is set. Finally, (^(s) _(i+1,desired),        b_(i+1,desired)) is chosen as follows:        S _(i+1,desired) =s, such that s·b _(s,i+1,desired)=max(s·b        _(s,i+1,desired))        b _(i+1,desired) =b _(s,i+1,desired), with s=s ^(i+1,desired)        With regard to the Maximum Squared Error Algorithm, instead of        computing Z_(1b,i), Z_(2b,i), Z_(b.i), M_(b,i), and h_(b,i), for        a single symbol rate s as described earlier, Z_(1s,b,i),        Z_(2,s,b,i), Z_(s,b,i), M_(s,b,i,), and h_(s,b,i), is computed        for all s in S. The primary difference in these computations is        that the following substitutions for E_(hdr,i) and E_(pld,i) are        made:

E_(hdr, s, i) ≡ E_(hdr, i) ⋅ λ_(s, i)${E_{{pld},s,i} \equiv {E_{{pld},i} \cdot \frac{\lambda_{s,i}}{\lambda_{s_{1},i}}}}\mspace{31mu}$Furthermore, the definition of “observable” is expanded, making thefollowing substitution for the condition b≧b_(i):

$d_{\min,b}^{2} \leq {d_{\min,b_{i}}^{2} \cdot \frac{\lambda_{s,i}}{\lambda_{s_{i},i}}}$Note that this condition reduces to b≧b_(i) when s=s_(i).b_(s,i+1,desired) is chosen as described earlier for b_(i+1,desired),but independently for each symbol rate s is S, using Z_(1s,b,i),Z_(2s,b,i), Z_(s,b,i), M_(s,b,i), and h_(s,b,i), in place of Z_(1b,i),Z_(2b,i), Z_(b,i), M_(b,i), and h_(b,i). Additional hold counts H_(s,i),for each symbol rate s in S are now updated:

$H_{s,i} = \left\{ {{\begin{matrix}0 & {{{where}\mspace{14mu} M_{s,\beta_{s},i}} \geq {\left( {1 - \Delta_{benefit}} \right) \cdot M_{\sigma,\beta_{\sigma},i}}} \\{H_{s,{i - 1}} + 1} & {{{where}\mspace{14mu} M_{s,\beta_{s},i}} < {\left( {1 - \Delta_{benefit}} \right) \cdot M_{\sigma,\beta_{\sigma},i}}}\end{matrix}{where}\text{:}\beta_{s}} \equiv {b_{s,{i + 1},{desired}}\sigma} \equiv s_{i,{desired}}} \right.$Finally, (S_(i+1,desired), b_(i+1desired)) is chosen as follows:

${s_{{i + 1},{desired}} = s},{{{such}\mspace{14mu}{that}\mspace{14mu} M_{s,\beta_{s},i}} = {\min\limits_{s \in S}\left( M_{s,\beta_{s},i} \right)}},{{{subject}\mspace{14mu}{to}\mspace{14mu} H_{s,i}} \geq h_{\min}}$b_(i + 1, desired) = b_(s, i + 1, desired), with  s = s_(i + 1, desired)In accordance with the present invention, both the Mean Squared Errorand Maximum Squared Error algorithms are likely to perform well in thepresence of quasi-static channels and white noise. Since the MeanSquared Error Algorithm assumes quasi-static white Gaussian noise, itperforms well in the presence of Time Invariant White Noise. Performanceis likely to be poor in the presence of Time Varying White Noise, sincethe noise level varies on the same time scale in which it's measured,leading to high variance in the estimate SNR _(i), and high variabilityin the chosen rate.Since it discards packets with payload errors whencomputing the estimate SNR _(i), it also performs well in the presenceof high level impulse noise, ignoring measurements from packets erroredby impulse events, and choosing the highest rate sustainable given thenoise floor. This reduces the likelihood with which an individual packetis exposed to an impulse event. However, performance is more ambiguouswith moderate level impulse noise, since moderate level impulse eventsmay occasionally bias packet measurements without causing errors. Thisleads to greater variance in the estimate SNR _(i), and greatervariability in the chosen rate. Performance is also likely to be poor inthe presence of Non-Gaussian Noise, if the algorithm assumes Gaussiannoise, and calculates the thresholds SNR_(min.b) accordingly. If thethreshold computation is generalized to use a more representative noisedistribution, performance could be improved. Since the Maximum SquaredError Algorithm can always observe the length independent Z_(1b,i) itperforms well in the presence of Time Invariant White Noise. Performanceis likely to be good in the presence of Time Varying White Noise, sinceboth Z_(1b,i) and Z_(2b,i) directly measure the likelihood of high noiselevels, even as the noise level varies rapidly across a range of noiselevels. The length independent measurement Z_(1b,i) is largelyunaffected by impulse events, since it is collected from headers whichoccupy little time on the medium. However, the length-dependentmeasurement Z_(2b,i) is directly influenced by error causing impulseevents. Together, the aggregate success rate estimate, Z_(b,i), and thetime on medium metric, M_(b,i), weigh the benefits of reducing time onmedium and the exposure to impulse events, against the cost of beingmore susceptible to them when they do occur. Thus, the Maximum SquaredError Algorithm performs well in the presence of both high and moderatelevel impulse events. Unfortunately, due to the problem ofunobservability, Z_(2b,i) can become inaccurate under certainconditions. These cases are addressed by the burst error limitingmechanism introduced earlier. The Maximum Squared Error Algorithmperforms well, since it measures packet errors directly without assuminga particular noise distribution.

Split Winding Transformer for Modem Transceiver S/N Optimization

Referring back to the NID analog front end shown in FIG. 4 b and aportion thereof shown in FIG. 69, in accordance with the presentinvention, a split winding transformer with turns ratios optimized formaximum transceiver signal to noise ratio is provided. More broadly, asplit winding transformer useful in a modem application is provided. Thetransmitter output signal level for typical modems is nominally fixedwithin some guardband of the FCC or other regulatory agency power limit.The signal level at the receiver input, however, is highly variabledepending on the channel attenuation in the path from a remotetransmitter. Consequently, the ideal line isolation transformer turnsratio from the transmitter output to the line of wt:1 is not optimal forthe receiver. At a modest additional cost of an additional transformerwinding, the turns ratios for the transmitter and receiver can be setindependently, while still allowing for hybrid echo cancellation. Sincethe receiver input signal will usually be less than the transmitteroutput signal, the optimal turns ratio is wt:1 from input to line, wherewr>wt. This step-up from the transmitted signal provides a “noiseless”gain that enhances the achievable receiver S/N ratio. The maximumseparation between wt and wr is limited by the reduction in couplingbetween transmit and receive windings that occurs for large differences.This introduces phase shift that compromises the effectiveness of commonecho cancellation schemes. Practical numbers for the wr:wt ratio arefrom 1 to 4. Prior art voiceband or ADSL modems do not take advantage ofthis technique. In the case of ADSL, the situation is particularlyegregious in that it is common to use step up transformers from modem toline side in order to boost the transmitted signals up to levelsrequired for long distance communication. This means there is actuallyattenuation of received signals. As can be seen in FIG. 69,filter/transformer/protection components 445, typically includingfilter/protection components 451 and transformer 453, is coupled in theTransmit and Receiver paths 495 and provides, for example, TX path 455and RX path 457 to be coupled to TIP and RING of Phoneline RJ11connector 450 through transformer 453. Wt:1 is transmit winding ratio;Wr:1 is the receive winding ratio; with the reference point, the twistedpair line, being 1. Transformer 453 couples the TIP line to the TXsignal path from electronic hybrid 440 via wt:1 windings. Transformer453 likewise couples the RING line to RX signal path via wr:1 windings.Therefore, in accordance with the present invention, a small signal onthe line being received can be stepped up, while on the transmit side onthe other hand, a stepping down can occur. Therefore in accordance withthe present invention a Wr of 2 provides a 1 to 2 step up, while on thetransmit side a Wt is 2/3 would, in essence provide a ratio of 3 to 1between the transmit and receive transformer windings. As set forth inFIG. 69 common core 459 is provided with three windings thereon, namelytip/ring winding 461 a, transmit side winding 461 b and receive sidewinding 461 c. The transformer is thereby optimizable to provide thebest signal to noise ratio for the transceiver.

Transmit Off Switch for Modem Receiver Noise Reduction

Referring again back to FIG. 4B, in accordance with the presentinvention, a method for eliminating or reducing the coupling oftransmitter noise into the receiver of a modem during periods when notransmission is occurring is provided. A modem operating in half-duplexmode typically leaves the transmitter connected full-time to the hybridand transformer devices performing 4-wire to 2-wire conversion frommodem to line, even though it is not active while a signal is beingreceived. From a signal perspective, this has no consequence. However,the noise contribution from the transmitter output to the receiver inputcan be significant in a low-power signal environment. The addition ofsimple switch 435 (e.g., a two transistor transmission gate in CMOStechnology) between the output of the transmitter (e.g., filter 430) andhybrid 440 reduces noise injected at the receiver input and thereforesubstantially improve receiver S/N ratio. Activation of the switch canbe incorporated into an automatic gain control loop with the minimumgain control setting causing the switch to turn off. Alternatively, aspecific gain control code can be assigned to activate the switch, whichcan then be turned off (disabled) and on (enabled) in a directed manner.

As can be seen in the typical NID depicted in FIG. 4B, electronic hybrid440 feeds signal from the transmitter back into the receiver. VGA 470has two pairs of inputs, one fed back from the transmitter, the other areceive input from line 106. Any signal coming out of the transmittercauses a self-echo path (e.g., through the transformer depicted in FIG.69) into the receiver that should be suppressed, such that the receiverdoes not get confused as to whether such self-echo is a signal comingfrom line 106. Noise also can get injected into the receiver from thetransmitting side, even during times when there is no transmitting,since the electronics components in the transmitting path can contributenoise, even when idle.

Therefore, in accordance with the present invention, when thetransmitter is not transmitting, transmit-off switch 435 provided in thetransmitting path, is switched off thereby blocking noise from gettinginjected back into the receive path which would deteriorate receiverperformance. As can be seen in FIG. 4B, in the preferred embodiment theswitch is located proximate to the end of the transmit path, i.e., justbefore combined electronic hybrid 440.

Voice Implementation Aspects

Given the Homenetworking implementation aspects of the present inventiondescribed above, we now turn to voice implementation aspects associatedtherewith which include concepts involving Voice over IP (VoIP) packetlatency on Homenetworking LANs, timing synchronization, timingsynchronization circuit, VoIP Head of Line (HOL) blocking solutionimplementation requirements, and collision signal slot assignment.

The desire to create quality VoIP service springs from an initiativewith several goals, including the reduction of the cost of maintainingcarrier networks by switching from a circuit-switched to apacket-switched model and by allowing voice and data to share a commoninfrastructure. Existing cable providers see this evolutionary step asan opportunity to provide new voice services in an effort to expandtheir core businesses. Existing voice carriers recognize the threat ofpotential new voice service providers and wish to merge theirvoice-networks with data networks in order to remain competitive.Consumers will expect these changes to result in the delivery of atleast an equivalent of their current voice service at a reduced cost. Inorder to satisfy the demands of all interested parties, the preferredoutcome of the VoIP initiative is to provide voice calls of a qualitywhich is at least equal to that delivered by today's circuit-switchednetworks. The quality of a voice call is affected by at least twometrics: (1) audio fidelity and (2) audio delay characteristics. Audiofidelity of packetized voice is affected by several parameters. Amongthem are: (1)Choice of codec—in general, the lower the compression ratein the codec, the higher the mean opinion score (MOS) of the resultingplayback, and the better the perceived speech quality; and (2) Lostsamples due to congestion and transmission errors and excessive pathlatency—lost samples result in distorted speech and other audibleartifacts, as well as poor throughput for voice-band data equipment suchas FAX.

Audio delay in a VoIP system is determined by the summation of theindividual delays that occur within the total communication path for thecall. The path includes codec delay, packetization delay, LAN queuingand transmission delay, IP network queuing delay, processing andtransmission delays, far-end LAN delay and finally, de-packetizationdelay. When this total delay exceeds about 150 msec one-way, theperceived response time of the party at the other end tends to exceedthe normally expected human response time. As a result, speakers tend tobecome impatient and repeat themselves and inadvertently interrupt theother speaker. The result is general annoyance, confusion andfrustration on the part of both speakers. Anyone having made aninternational call through long transmission links may have experiencedsuch behavior. Employing simplex-channel handshaking (i.e., saying“roger” to indicate when one is finished speaking and awaiting aresponse) is not an acceptable solution.

Referring back to FIGS. 1 b and 1 d, the cable network can be a DOCSISnetwork with completely structured time periods. The Cable ModemTermination System is essentially a “master” on the cable network anddictates to the various cable modems (“slaves”) in the homes exactlywhose turn it is to access the cable system network and when, therebyavoiding contention. The operation of the Cable Modem Termination systemin conjunction with Cable Modem(s) is described in the aforementionedpatent application incorporated herein by reference. However, the sharedHPNA network described above does not have a centralized master who isdictating timing on the network. The shared home networking uses therandomization process to sort out the accessing. You can specificallytime the highest priority traffic (e.g., voice) to minimize the time ittakes for a voice packet to make it from one end of the cable to theother. This is necessary to preserve the quality of voice, otherwisethere are difficulties in communicating over the telephone. When thevoice packet enters the home it is necessary to transfer it from thegateway to the handset, and deal with the other devices operating on thehome network (e.g., PCs, printers, etc.), or other handsets for thatmatter which may have multiple calls operating at the same time.

A complete VoIP system includes the IP backbone, service providerhead-end equipment, a local delivery network and finally, a homedistribution network. Referring briefly to FIG. 70, there is shown adiagram of the network including a plurality of POTS phones 2016 a–2016c (with an analog connection to respective converter devices 2017 a–2017c which convert the analog calls into samples and packetizes them forsending over the network) and an HPNA VoIP phone 2016 d (a phone with anHPNA transceiver built into it). The converter device needs to operateat the precise 8 kHz clock which is described through the protocol inaccordance with the present invention. In addition, if samples are beingcontinually taken at 8 kHz, an issue arises as to at what point is a setof those samples taken called a packet and started being sent. This willrelate to the cable modem system network which has a deterministictransmission algorithm such that the CMTS (head end) 2020 will dictateto all of the nodes and process grants associated with requests from thegroups of homes having the cable modems. The head end sends a singlepacket out that describes which home is next in the rotation in terms ofbeing able to deliver a packet. The gateway 2018 sets up with the headend the fact it would like to get the DOCSIS unsolicited grants (at 10msec intervals). The headend assigned the unsolicted grants. However,the devices on the HPNA LAN are not privy to the information that isbeing transferred back and forth on the DOCSIS network. As such, thedevices are not aware as to when the grants occur. A packetsynchronization mechanism is therefore needed to determine when thegrants occur on the network and then tell the sample framing to set upthe set of samples so that they can be sent on the HPNA network andarrive just in time for the grant received by the home gateway. Inessence, the grant information is obtained from the cable modem,delivered on the HPNA LAN network to the appropriate node (each nodepossibly having a grant on the DOCSIS network having a different timeplacement). Once the individual node has its individual information, itthen takes a stream of samples from a telephone (e.g., every 125 usecanother sample is obtained, wherein 80×125 usec is 10 msec). A set of 80contiguous samples is needed to be picked at some point in time, and theconverter node needs to understand when the grant needs to have thepacket at the cable modem, and then some time back from there the 80sample packet needs to be delivered to the cable modem via the HPNA LANsuch that when it gets to the cable modem it is not too late for thegrant. If it arrives too late it will have to wait for the next grant,which is 10 msec away, adding another 10 msec of time it takes to getacross the entire network. Referring briefly to FIG. 71 an upstreamtransmission of a VoIP frame #N is seen. This indicates when the grantarrives. Another grant occurs with a 10 msec time between the grants.Therefore, what is needed is a determination such that the converterframes up the sample far enough back in time that it will be sent alongthe HPNA network in time to reach the grant. The present inventionprovides such a determination.

VoIP Packet Latency on Home LANS

It is therefore important to minimize the latency of packets travelingover the network. It also important to minimize the variance in thetiming of the samples taken at one end and the play out at the otherend. The present invention achieves both of these objectives. Inaccordance therewith, frame slip is involved with the difference betweenthe sampling end and the playout end in terms of the clock used at thehead end and refers to an individual sample that was taken at an 8 kHzrate. In this context, frame slip means that, if samples are taken atslightly higher than at 8 kHz rate and played out at the 8 kHz rate,eventually there will be more samples than time to play them out,necessitating a drop in samples in order to catch up, such droppagebeing termed frame slip, a frame being one sample at 8 kHz. The idea isto minimize the number of frame slips, preferably a tolerable amountbeing 0.25 frame slips per minute. In order to accomplish this tolerableamount there must be some mechanism for conveying the clock informationacross the network so that the same 8kHz clock is being used, ratherthan one that is a little faster, or a little slower. The presentinvention provides such a mechanism.

With regard to path latency and the jitter in that latency, it is notdesirable to have packets assembled on one end of the network andleisurely delivered at the other end. The generally accepted number forthe longest time that can be taken in delaying a packet when it isdelivered across the entire network (i.e., from the user home telephone,through the home network, up the cable, through the internet, andsimilarly back down the other end to the other user home telephone) is150 msec. Anything beyond that latency creates stilted conversation. Inorder to keep the latency amount down, several issues must be dealt within a shared network. If standard Ethernet is used the variance inlatency that is caused by the standard backoff algorithm will costdearly in what can be tolerated for voice transmission. In the HPNA V2protocol, which also implements a randomized backoff algorithm, it isdifferent than standard Ethernet in that winners do not get rewarded forhaving won. A winner has to wait out until all the losers of thecontention round have each gotten their turn before the winner can goback with a new transmission. This has the effect of greatly reducingthe latency on the collision resolution and provides a good basis forvoice transmission.

Components to latency can include the possibility that someone else isusing the network when the network is desired to be used and HPNA hasprioritization built into it, with voice traffic being given the highestpriority. However, such does not stop someone else being on the networkfirst, and the latter needs to wait its turn to send out a packet eventhough it has highest priority. When the prior sender is done, thelatter will have highest priority to the idle network. A collision mayalso occur with other nodes having highest priority traffic. Thecollision resolution in the worst case can take up to 2.7 msec with aspecific probability of resolution. If the probability is extended tohigher and higher numbers it takes longer and longer. This 2.7 msecnumber is taken as the target number for the performance currentlygenerally accepted in the industry. Another general requirement in theindustry is that four total telephones all doing the same thing at thesame time using the same network to the same gateway must be supported.Another assumption made is that the transmission is ruined by an errorand it must be all done again. The total time it takes being 11.8 msec,the worst case. In the other direction, the gateway has a queue ofpackets (transmitting data for four nodes) in the downstream direction,but it only represents one point of attachment to the network. If thegateway's traffic collides with all four coming up, a five way collisionoccurs, and if the gateway is the last one to win in that collision,then the gateway has to wait for all of the other upstream nodes tofinish before it proceeds. Everybody gets knocked out due to noiseagain, and then all that gets out is one node's voice packet. The otherthree have to wait to get out in the downstream direction. The total,with there being 11.8 msec on one side of the call and 14.9 msec on theother side of the call, is 26.7 msec. With 150 msec to make the completeend to end trip, then 114.3 msec is left for the rest of the trip. Inaccordance with the present invention improvements have been made toimprove upon the latency and synchronize the clock.

In addition to the voice quality issues raised above, anyhome-networking solution for VoIP call distribution must support areasonable number of simultaneous calls and must be made available at areasonable cost. Each of these requirements represent constraints on theeffort to meet the overall call quality objective. The qualitativeperformance objectives are summarized by the more specific technicalrequirements given for the entire system as follows:

-   1. The frame-slip rate (voice-sample loss rate) of the entire VoIP    path should not exceed 0.25 slips per minute.-   2. The end-to-end path latency for any voice call should not exceed    150 msec—the home LAN portion of this overall requirement is 10    msec.-   3. The home LAN should support 4 simultaneous VoIP calls in order to    provide an adequate and competitive level of service to the end    user.-   4. Cost of installation must be reasonable.    As mentioned above, a frame slip is the loss of a single 8kHz    audio-band sample. By minimizing frame slips, packet loss is    minimized, and speech quality is preserved. In addition to problems    encountered with speech, voice-band data traffic suffers severe    throughput degradation when even a small number of codec samples are    lost. Because each VoIP packet that travels on the home network    usually includes at least 80 voice samples, the loss of a single    VoIP packet will cause 80 frame slips. Only one such loss is allowed    per 320 minutes if the 0.25 frame slip per minute goal is to be    achieved. This places a very strict requirement on the home network    operation with regard to packet loss. Packet loss in a home network    could be due to any of several factors, including:-   1. Losses incurred due to unrecovered bit errors in the transmitted    message.-   2. Timing mismatch between the 8 kHz codec sampling at the    transmitter and the timing slots that exist on a synchronous network    (some VoIP calls may traverse a synchronous network for at least a    portion of the route) which ultimately leads to buffer overruns or    underruns.-   3. Late packet arrival—packets which arrive too late will be    discarded, and are effectively, lost. For example, at the home    LAN/WAN interface, late-arriving upstream VoIP packets will be    discarded, since they cannot be expected to arrive on time anywhere    else along the path once they are late at the WAN interface.-   4. Congestion loss—some LAN protocols may discard queued packets due    to congestion or other failures, e.g., excessive collisions in an    IEEE 802.3 network causes a packet to be dropped.

As also mentioned above, 150 msec is allotted for the total VoIP callpath. The portion of the call path latency allocated for home networkdistribution is 10 msec. Assuming that both ends of the call areterminated in a home LAN, then the packet delivery latency allocated tothe home network at either end of the call is roughly 5 msec. It shouldbe noted that any VoIP packet which arrives later than the allowed 5msec latency will effectively be lost, even though the packet may havearrived without errors. A late packet cannot be used in speechreconstruction. Therefore, the ability of a network to meet the requireddelivery latency objective not only affects the audio delay figure, butit also directly affects the rate of frame slips. Exploring thisrelationship further reveals the following limit on latency: (1) Thenetwork can tolerate the loss of a single packet in 320 minutes; (2) Onedirection of a VoIP connection will pass 320*60/10 msec=1.92E6 packetsin 320 minutes; (3) Therefore, the allowed rate of lost packets is1/1.92E6=0.52 packets per million. For the latency requirement, thismeans that the 5 msec one-way latency number must be met 99.99995% ofthe time. Most LAN protocols proposing to fill the home networking spacewill be incapable of insuring that such a figure is met, becausetraditional asynchronous networks provide highly variable latency. Somenetworks might be able to achieve a 5 msec average latency, but thejitter in the latency figure would cause some significant percentage offrames (i.e. >>0.0001%) to be delivered with a latency above the 5 msecnumber. This level of performance will not conform to the objectives ofa high-quality VoIP system.

Standard Ethernet protocol might be one choice for a home LAN, if itwere to be adapted to a home-friendly medium, such as a phone line or awireless carrier. The HPNA V1 protocol was built in just this fashion.HPNA V1 protocol is the IEEE 802.3 MAC protocol operating on home phonelines with a novel PHY implementation. However, the IEEE 802.3protocol's collision resolution algorithm delivers very unsatisfactorylatency performance when the aggregate network load is moderate to high.One of the most often-cited problems arising from high offered load onthe IEEE 802.3 protocol is the network capture effect, where competingstations take turns at virtual dominance over the network for relativelylong periods of time. Overall, the network behavior is fair, but forshort periods of time, the network is very unfair. The duration of theunfair access times is much greater than the required 5 msec deliverylatency for a single VoIP packet. A question arises as to whether VoIPtraffic can create high enough network loads to cause the networkcapture effect to appear. If the network is to also be shared for datatraffic, then the answer is very clear: A high load can be attained withthe introduction of just one network session which attempts to transfera medium-sized file or web-page. Such a file transfer will cause aninstantaneous load increase which is sufficient to push the 99.99994%confidence envelope for jitter well beyond the 5 msec number. Thefollowing explanation gives an example: The IEEE 802.3 network captureeffect is the result of the allowance for the winner of a collisionresolution to return to the network with the next frame in itstransmission queue, with a built-in advantage over the previous loserfor the next round of collision resolution. Because of the built-inadvantage, the loser has a relatively high chance of becoming locked outof the network for an extended period of time roughly averaging:3.5×2^10×51.2 usec (network slot time)=184 msec for a 10 Mbit network.Following this average capture time, the loser would discard thecurrent-transmit frame due to excessive retry failure. During thenetwork capture event, the losing station would have generated anadditional 18 VoIP packets. Each of these would in turn be discarded atthe LAN/WAN interface, because they would all arrive much past theirrespective 5 msec latency limits. This utter failure puts the IEEE 802.3protocol well outside of the required performance objective. The HPNA V1protocol utilizes the IEEE 802.3 MAC function and it specifies a slottime for backoff purposes of about 300 usec. For HPNA V1 systems, thisresults in an average network capture event duration of over 1 second.Clearly, a more sophisticated protocol was required. Within the IEEE802.3 MAC protocol, some provision can be adopted to reduce the networkcapture effect, such as applying traffic scheduling to minimize theduration of capture events. Unless the scheduling operation is performedin the device driver which has real time knowledge of the MAC TX queueand of the overall network load, the result will be ineffective. Evenwith effective scheduling, there can be no guarantee that the nextwinner is a node with VoIP traffic instead of the latest download fromZDNET. Average latency can be reduced, but extremes will still be beyondthe necessary maximums, and the most latency-sensitive traffic is notgiven any priority. A better alternative is the one brought forward byHPNA V2, as described above.

The HPNA V2 protocol solves the problem of network capture by employinga simple, yet tightly-bounded collision resolution mechanism which doesnot rely on a central arbiter to resolve network sharing issues, knownas DFPQ, as described above. The key to the protocol is the employmentof a RX based collision detection scheme that allows all network nodesto maintain consistent knowledge of the network condition. Based on thenetwork condition, each node can determine whether it may take its turnto transmit a packet. Once a node has transmitted, it must wait untilall other nodes wishing to transmit have also taken a turn before it canget a new turn. As multiple nodes contend for the same opportunity totransmit, they perform a randomization step which serves to define atransmission ordering among the competing nodes. Because the orderingdecision is random and decentralized, there is some non-zero tail to thetime it takes to resolve a non-colliding order, but the tail is verysmall in comparison to the behavior of more traditional collisionresolution protocols. By utilizing DFPQ collision resolution, thenetwork capture effect is completely avoided. Removing the networkcapture effect is not enough to completely solve the problem ofdelivering guarantees of limited latency. Because some competing nodesmay have relatively time-insensitive packets in their queues (such asfile transfer data or web-page data), the latency of delivery for thetime-sensitive VoIP packets could be greatly altered by the varying loadof time-insensitive packets. Worse yet, if there were a large number ofnon-VoIP nodes in the network, and each was given a turn over the VoIPnodes, then the aggregate of non-VoIP traffic could add up to more than5 msec. This would again, cause the desired latency bound to beviolated. However, DFPQ again provides a solution. This time, thesolution is to allow for multiple instances of the MAC protocol tooperate in parallel, with one instance for each of 8 different prioritylevels. Each MAC instance (priority level) operates with a successivelylonger inter-frame space (IFS). Packets from lower levels of priorityare only allowed access to the network when no packets exist at higherlevels. This mechanism prevents time-insensitive traffic from affectingthe delivery latency of packets from higher levels of priority. VoIPtraffic is assigned to the highest level of priority because it has thetightest performance targets of all network traffic. Therefore, VoIPtraffic always gets to “cut in line”, ahead of lower priority traffic.

In order to take full advantage of the multiple priority levels of theHPNA V2 protocol, a MAC controller includes multiple transmissionqueues. If only one transmission queue exists, then it is possible forthe device driver to load a low-priority frame into the queue, notknowing that a short time later, a high priority frame may need to bequeued. When the high priority frame does arrive at the device driver,this frame must wait until the low-priority frame is transmitted by theMAC. The HPNA V2 protocol employs absolute priority, such that lowerpriority frames are completely blocked from network access until allhigher priority frames from all nodes of the network have beentransmitted. Therefore, a high priority frame waiting in a transmissionqueue behind a low-priority frame will potentially wait for a longperiod of time before being transmitted. Because the longest allowedframe transmission time is 3.2 msec, the transmission of only 1higher-priority frame can cause the blocked frame to be late. Referringto FIGS. 72 a and 72 b, all of the frames with higher priority fromNodes B, C and D will be transmitted before the PRI=0 frame from Node A.The blocking action of the PRI=0 frame causes the highest priority frameof all queued frames (the PRI=7 VoIP frame at Node A) to wait behind 8frame of lower priority. This behavior is not acceptable and causeslatency for VoIP frames beyond the 5 msec limit. The solution to theproblem is to allow multiple transmit queues at the MAC, such that thePRI=7queue is serviced ahead of others. In accordance with the presentinvention, a single physical queue is used, with queue managementcontrol that allows re-ordering of previously queued frames to allow thehighest priority frame to appear at the head of the queue. Now the VoIPPRI=7 frame will be the first to access the network, ahead of all otherframes on the network. Further details as to VoIP HOL Blockingimplementation requirements are set forth hereinbelow.

In qualitative terms, the latency of delivery of frames using DFPQ isgreatly improved over previous methods. However, as defined in the HPNAV2 characteristics, the latency delivered by DFPQ does not quite meetthe performance goal of 5 msec, so a few additional features have beendefined to achieve the final desired result.

Referring now to FIG. 73, which shows aspects of FIG. 70 in more detail,there is seen stratum3 reference clock 2044, which is a very highaccuracy clock which is delivered to headend 2020 at the serviceprovider and delivered down the cable to cable modem 2012. Cable modemclock 2013 is provided to HPNA transceiver 2015 at the cable modemgateway. The HPNA transceiver then has a mechanism to deliver the clockto a representatively depicted HPNA-pots Converter 2017 a. Voice codec2050 takes the analog signal 2019 and converts it into digital samples.The digital samples get delivered to HPNA MAC 2015, turned into a packetand gets delivered up to the network. In delivering the clockinformation over the DOCSIS system it was through a special packet whichgot captured by a special circuit. However, the CMTS and CM clockcoordination could rely on the latency of delivery because the DOCSISnetwork provides a mechanism for determining how far the CM is on thewire from the CMTS. This is done for every node. When the CM receives aclock stamp it adjusts it according to its distance from the CMTS. Also,the CMTS knows exactly when the clock is relative to the time when itactually sends the signal out onto the network. The HPNA network, beinga shared network, does not have this adjustment. Therefore, inaccordance with the present invention, the clock is taken from the cablemodem to the transceiver and the transceiver takes the clock informationand runs a clock inside based upon the clock information from the cablemodem. At an appropriate time, a special frame is created that capturesthat time and delivers it across HPNA V2 network 2014, the transceivergateway being considered a “master”, with each HPNA−VoIP convertertransceiver being considered a “slave”, to all of the slaves. The slavesdo not correlate their clocks directly. All they do is discover thedifference between their clock and the other (master) clock. The slavestake their own local timestamp based on their local version of the clockand look at the time. They do this over multiple periods to find thedrift between their clock and master clock. Once they do this they cantake the information and use it to adjust the speed of their clock witha DPLL circuit.

In order to explain the new features in more detail, it is illustrativeto employ an example. Referring again to FIGS. 70 and 73, there areseveral components to the latency in the HPNA LAN portion 2010 of theVoIP packet transmission path. CM CPE 2012 is a cable modem device. Inthe example network, HPNA LAN 2014 connects 4 VoIP terminals 2016 a,2016 b, 2016 c, 2016 d to home gateway 2018. POTS phones 2016 a, 2016 band 2016 c are connected to HPNA LAN 2014 via converters 2017 a, 2017 b,2017 c, respectively. VoIP terminal 2016 d can have a converter builttherewithin and be thereby connected directly to HPNA LAN 2014. Homegateway 2018 interconnects with CMTS 2020 over an HFC line 2021 usingDOCSIS protocol, which can be located in, for example, telephone companyCentral Office 2024. CMTS 2020 can service the PSTN network (“cloud”)2026 through PSTN switch 2028. CMTS 2020 can also service the Internetnetwork (“IP Cloud”) 2030 via IP through IP router 2032. CM 2012 is alsocoupled to HPNA LAN 2014 through HPNA MAC 2015. The support of 4simultaneous VoIP connections was given as the required level of supportin the original set of performance goals. Within the HPNA LAN portion ofthe system, there are several components which contribute to the totallatency of the LAN. These components and their individual contributionsare listed in the table set forth in FIG. 74. Note that upstream trafficexperiences a shorter latency than downstream traffic. This is due tothe fact that the downstream traffic originates from a single MAC source(i.e. the HPNA MAC within the Cable Modem device) and therefore mighthave to wait for all nodes to transmit before each of its third andsubsequent transmissions. The latency components in FIG. 74 are definedas follows:

-   Access Delay This is the maximum time that a VoIP node will have to    wait if a VoIP frame is queued just as a maximum-length transmission    begins at the lowest possible HPNA V2 transmission rate (=4    Mbit/sec). There is no network preemption if a frame has already    begun transmission.-   Collision Resolution This is the overhead needed to resolve a    five-way collision among 5 VoIP transmitters (4 handsets upstream    frames+one CM downstream frame) to the level of uncertainty of 10E−6    or 10E−1, depending upon the column heading. The 10E−6 value is    based on the latency of 13 collision resolution cycles. The overhead    includes the time needed for collision events mixed with IFS times    and the collision resolution signaling function. Transmission of    packets from winning nodes is listed separately.-   3 Up, 1 Down This is the time required to transmit 3 VoIP packets in    the upstream direction, plus one VoIP frame in the downstream    direction. This activity would normally occur interspersed with the    Collision Resolution overhead time.-   Last Up This is the time required to transmit the last VoIP packet    in the upstream direction.-   3 down This is the time required to transmit the last three VoIP    packets in the downstream direction.    The 3 Up, 1 Down, collision resolution, and 3 down entries are    repeated on the HPNA LAN in order to bring the delivery reliability    of all VoIP packets to an acceptable level. The different columns    indicate the differing probabilities of the event combinations. Head    of Line blocking is assumed to be solved in the table. If not    previously solved, the contribution from this component would be    potentially very large. The different columns represent different    cases of probability. The 10E−6 column includes the time needed to    resolve a t-way collision to 10E−6 probability that all stations    have had a chance to transmit. This column also assumes that the    traffic occurs in a home where the network transmission rates for    all 5 nodes have been negotiated to 4 Mbit/sec. In the 91% column,    the collision resolution is still 10E−6 probability, but the    transmission rates for all nodes has been improved to 10 Mbit/sec.    The 90% probability case assumes collision resolution time with 90%    probability and network rates of 10 Mbit/sec. As can be seen in the    table, the total latency for a one way path exceeds the 10 msec    budget allocated to the HPNA LAN portion of the complete path.    Several additional features are defined in order to bring the home    LAN latency to a more suitable level.

The collision resolution mechanism of HPNA V2 can take more than 2 msecto resolve in the extreme case of probability when 5 nodes collide.Improved algorithms for reducing the collision resolution overhead havebeen described above. Analysis of these mechanisms demonstrates a vastimprovement in the time required for resolution of collisions using thenew methods. The FIG. 74 table shows a VoIP frame length of 160 bytesfor 80 samples. This translates to a 16-bit linear PCM coding of thevoice from the handset. (A 16-bit PCM code would include only 14-bits ofresolution, but the extra two bits are needed to allow alignment to byteboundaries.) If PCM μLaw is used instead as the codec, then the numberof payload bytes per packet is reduced from 160 to 80. This eliminatesanother 2.4 msec of delay from the downstream side and 1.5 msec from theupstream side. PCM-μLaw is the codec currently employed as the standardfor all US PSTN traffic. While the BER for the HPNA V2 network can bemanaged to very good values by employing a rate negotiation feature,there are still potential impulse noise events that can cause packetloss. The simplest recovery from such events is to unconditionallyprovide two separate copies of every packet. Other mechanisms to protectagainst impulse-noise induced frame loss either are not effectiveagainst impulse noise (e.g. FEC) or cause excessive additional latency(e.g. low-level ARQ). Therefore, the table shows that every VoIP packetis being transmitted twice in order to protect against frame loss due toimpulse events. This brings the HPNA V2 LAN BER-induced frame slipperformance to the required levels, but it does add to the latency ofthe system. Delivery latency must be small for voice (<5 msec) becausethe overall path latency in one direction should not exceed 150 msec.Frame slips must be rare because they cause a resynchronizationprocedure in voice-band data connections which causes a complete halt ofuseful information transfer, thereby drastically reducing throughput.Voice traffic is not as sensitive to frame slips as voice-band data.Voice-band data is not as sensitive to latency as voice traffic.Therefore, if true voice traffic can be separated from pure voice-banddata traffic, then the voice-band traffic can be sent at a lowerpriority, where the latency of delivery will not be as tightly bounded.This reduces the constraint on HPNA LAN performance, because the frameslip rate can probably be relaxed for the pure-voice calls, therebyeliminating the need for the double transmission of frames and reducingthe latency as a result. Using this method, voice packets which are lostdue to impulse events will not be recovered. Voice-band data packetswhich are lost due to impulse events are recovered using HPNA V2standard LARQ procedures (low level ARQ) as described above. Anotheroption is to allow the redundant transmission to occur withoutconcentration—i.e., use a very short IFS to transmit the complete frametwice in a row. This eliminates the second round of collisionresolution, trimming another 0.4 msec from each end of the path. Thechart set forth in FIG. 75 shows the latency when PCMμ coding is usedinstead of 14-bit linear PCM, and the redundant VoIP packettransmissions are dropped. A mechanism for improving collisionresolution time is also included. These numbers are very close to therequired latency performance of the VoIP end-to-end system.

An additional system latency component exists. This component is due tothe lack of coordination of the framing of voice samples at the handsetsuch that a VoIP packet will be sent on the LAN and arrive just in timeto utilize the next available upstream transmission slot on the WAN. Ifsuch coordination is not performed, as much as an additional 10 msec oflatency can be added to the upstream path. As an example, the DOCSISnetwork for cable modems allocates a fixed amount of bandwidth to theupstream portion of each voice call. The bandwidth is allocated inperiodic intervals which match the framing rate of the VoIP frames. Forexample, a call utilizing a 10 msec VoIP packet rate will receive aDOCSIS upstream bandwidth allocation allowing it to transmit one frameevery 10 msec. The initial timing of the upstream transmissionopportunities is random, relative to the potential framing of a set ofsamples at the handset. If the handset creates a VoIP frame and sends itto the cable modem, with the arrival just missing an upstreamtransmission slot, then the VoIP frame will have to wait 10 msec to beforwarded to the central office. Previous to arriving at the cablemodem, the VoIP frame already experienced the 5 msec allotted LANlatency. Referring back to FIG. 71, this timing relationship isillustrated, where the packet arrival at the cable modem is too late forthe current upstream transmission slot, resulting in an additional 10msec of latency for delivery of all VoIP frames in this stream. As canbe seen in the illustration, the total latency from packetization at thehandset to delivery on the DOCSIS network for this example is about 15msec. This is well beyond the 5 msec target. If the packetization at thehandset can be synchronized to accommodate the HPNA LAN delivery latencyand the cable modem processing delay such that the VoIP frame is readyfor transmission on the next upstream slot, then the additional 10 mseclatency penalty can be avoided. The mechanism for coordinating thehandset framing is accomplished through a protocol to communicate theupstream slot timing from the cable modem to the handset. Suchcoordination is provided via a clock synchronization mechanism betweenthe cable modem and the handset. The clock synchronization mechanismincludes a timing circuit within the HPNA MAC controller and an HPNA LANprotocol for the exchange of timing information. Through the exchange oftiming information, the handset discovers when the next upstreamtransmission opportunity will occur. It assembles the initial VoIPpacket at such a time that accounting for HPNA LAN delivery latency andcable modem processing latency, the packet will arrive in time for thenext available upstream transmission slot.

Referring again to FIGS. 70 and 73, even if all VoIP packets from theHPNA-enabled handset are delivered error-free and on-time, a VoIP callmay still experience frame slips which are due to a mis-synchronizationof the handset A/D sampling clock and the reference clock of thesynchronous network which provides a portion of the transport for theVoIP packets. In the example thus far depicted in FIGS. 70 and 73, theDOCSIS network is synchronous, and so is the PSTN over which the callmay eventually be routed. In either case, if the VoIP packet samples arecollected at a rate which is slower or faster than the standard 8 kHzrate, then there will eventually be an accumulated sample deficit oroverage. The result will be that eventually, an entire VoIP packet willbe lost. Because of the 0.25 frame slip per minute requirement, the 8kHz sample clock at the handset must be accurate to within 0.52 ppm ofthe synchronous network clock's time reference. The synchronization ofthe handset sample clock to the synchronous network's reference clock isaccomplished through the same protocol that allowed the conveyance ofupstream slot timing information. The protocol employs a timestampgenerator in the cable modem and the handset. Cable modem 2012 alreadymaintains a clock in timing synchronization circuit 2040 which is lockedto the DOCSIS head-end timing synchronization circuit 2042 and in turnto reference clock 2044, because all DOCSIS network activity issynchronous to this reference clock. Cable modem 2012 makes thissynchronized clock available to HPNA MAC device 2015, which then uses itto drive timestamp circuit 2046. The HPNA MAC timestamp is deliveredfrom cable modem 2012 to handset devices 2017 a by way of HPNA LAN 2014.Handset MAC devices use these timestamps to synchronize their localclocks in their respective timestamp recovery circuit 2048. They inturn, provide an output clock which is used by the A/D sampling circuitin voice codec 2050. The HPNA timing information exchange messagesinclude absolute time references which are used to synchronize thehandset clock with stratum 3 timing reference 2044, and they includeupstream transmission slot information which allows VoIP packets to beframed and delivered at the most appropriate time.

Time Synchronization

Signaling frames and procedures are defined to permit timesynchronization between Home gateway 2018 and representative HPNA-POTSconverter 2017 a as depicted in FIG. 73. The time synchronizationprocedures enable two types of time synchronization: (1) The 8 kHzsample rate of the analog voice codec at the handset is synchronized toa reference clock at the Home gateway; and (2) The generation of encodedvoice packets at the HPNA-POTS converter is synchronized to the arrivalof the assigned upstream timeslot at the Home gateway from the digitalcarrier network, accounting for any processing delays or jitterintroduced by HPNA network access. In the DOCSIS/PacketCable system,this is the arrival of an upstream grant sync for the service flowallocated for the specific voice stream.

Referring to FIG. 73 in conjunction with FIG. 80, home gateway 2018implements a counter/timer that is sync-locked to the network stratumreference source. The HPNA MAC transmitter in the Home gatewayimplements a function to read and latch the value of the counter/timerinto Master Timestamp Register 3011 at the exact time of transmission ofa frame marked with the “Latch Timestamp” (LTS) descriptor bit. TheHPNA-POTS converter implements a counter/timer which is subdivided toderive the Codec clock. The HPNA MAC receiver in the HPNA-POTS converterimplements a function to read and latch the value of the counter/timerinto Receive Timestamp Register 3013 upon the receipt of a frame.Receive Timestamp Register 3013 is logically part of the receive statusword of each received frame. The timing information is conveyed to theHPNA-POTS converter via a pair of messages. The Home gatewayperiodically transmits a Timestamp Sync (TSM) frame with the LTSdescriptor set, then reads and transmits the latched Master Timestampregister value in a subsequent Timestamp Report (TRM) frame. TheHPNA-POTS converter reads and saves the Receive Timestamp registervalues of Timestamp Sync frames, and builds a database of correspondingReceive and Master timestamp pairs from the received TSM and TRM frames.The HPNA-POTS converter periodically calculates: frequencyerror=[(R₂−R₁)//M₂−M₁)]−1. The frequency error adjustment is thenapplied to the HPNA-POTS converter local codec clock.

The Home gateway implements a function to read and latch the value ofthe reference counter/timer into Grant Timestamp register 3030 upon theoccurrence of a selected timeslot grant sync signal from the upstreamnetwork (i.e. SID match and Grant sync). The Home gateway is aware ofthe mapping of upstream timeslot grant to specific HPNA-POTS converterand line ID. The HPNA-POTS converter implements a timer that generates alocal frame sync signal at the expected voice frame rate. This timer isderived from the local codec clock. The relative timing of the upstreamgrant sync signal is conveyed to the HPNA-POTS converter prior toenabling the voice encoder, but after the establishment of the upstreamservice flow. The timing offset is adjusted to account for internalprocessing cycles needed each by the Home gateway and the HPNA-POTSconverter, and allowing for worst case voice frame latency on the HPNAmedia. When the Home gateway needs to send the timeslot grant synctiming information, it will latch the grant timestamp value and adjustthe value to account for internal processing time to receive and forwardvoice frames to the upstream network interface. The adjusted granttimestamp is transmitted to the HPNA-POTS converter in a TimestampReport (TRM) frame. The HPNA-POTS converter calculates an absolute timeoffset from the difference in the Receive and Master timestamps, andcalculates a future local frame sync time as: Frame Sync=Granttimestamp+offset+voice frame period−latency; where latency=HPNA-POTSconverter internal processing time+worst case HPNA media transmitlatency. The method by which the Frame Sync adjustment is then appliedto the HPNA-POTS converter voice encoder is implementation-dependent.FIGS. 76 and 77(1)–77(2) depict the Timestamp Sync Frame format and theTimestamp Report Frame format, respectively. The Home gateway transmitstime synchronization frames (Timestamp Sync Message and Timestamp ReportMessage) on a periodic rate continuously. Frames are transmitted to thebroadcast MAC address using MAC priority level 6. Time sync messages arealways transmitted in pairs, according to the following procedure. TheHome gateway maintains a Time Sync timer and a sequence number counter,SeqNum. Upon expiry of the time sync timer, the Home gateway:(1)restartsthe Time Sync timer with period 1 second; (2) incrementsSeqNum=SeqNum+1; (3) formats a Timestamp Sync Message frame with thecurrent value of SeqNum; (4)marks the frame with the LTS=1 descriptorand (5) transmits the TSM frame. The Home gateway then: (1) reads thevalue of the Master Timestamp register; (2) formats a Timestamp ReportMessage frame with the current values of SeqNum and Master Timestamp,and (3) transmits the TRM frame. Upon the establishment orre-establishment of an upstream service flow for a media stream, theHome gateway: (1) obtains the grant timestamp for the service flow fromthe Grant Timestamp register; (2) adjusts the grant timestamp by a knownconstant equal to the internal processing time to receive and forward anupstream voice packet; (3) formats a Timestamp Report Message frame asabove, including the additional Grant Timestamp and associated Line IDand Call ID fields; and (4)transmits 3 copies the TRM frame. TRM framescontaining a Grant Timestamp are transmitted immediately (withoutwaiting for the Time Sync timer to expire). An HPNA-POTS converterderives clock and grant timing information from received Timestamp Syncand Timestamp Report message frames. Frames which are received with aMAC source address (SA field) that do not match the expected Homegateway are discarded. The HPNA-POTS converter maintains an informationbase of {SeqNum, Receive timestamp, Master timestamp} tuples. The mostrecent 2 tuples are retained; older tuples are discarded. Upon receiptof a Timestamp Sync Message frame, the HPNA-POTS converter reads theReceive Timestamp receive status word, and enters the {SeqNum, ReceiveTimestamp} tuple into its information base. Upon receipt of a TimestampReport Message frame, the HPNA-POTS converter: (1) locates the tupleassociated with the received sequence number, SeqNum, from itsinformation base; (2) enters the Master timestamp value in thecorresponding tuple in the information base; (3) calculates a codecclock frequency error: where frequencyerror=[R_(seqnum)−R_((seqnum−1)))/(M_(seqnum)−M_((seqnum−1)))]−1; and(4) adjusts the local clock frequency as necessary. When the HPNA-POTSconverter receives a Timestamp Report Message frame containing a GrantTimestamp, the HPNA-POTS converter: (1) examines the SeqNum field anddiscards the message if a duplicate received frame and takes no furtheraction; (2) examines the Line ID and Call ID field and discards themessage if no match to an existing voice call; (3) calculates the timedelta to the next local frame sync signal as follows: Frame synctime−Grant Timestamp+T_(offset)+VF−K_(cpu)−K_(HPNA); whereT_(offset)=Receive Timestamp−Master Timestamp (absolute time offset);K_(cpu)=a known constant equal to the HPNA-POTS converter internalprocessing time to prepare an upstream voice packet; K_(HPNA)=a knownconstant equal to the worst case HPNA media transmission delay; andVF=voice frame period; and (4) adjusts the local frame sync timing asnecessary.

HPNA VoIP Timing Synch Circuit

In accordance with the present invention a solution to the problem ofsynchronization of clocks between the Cable Modem (CM) and the handsetin a VoIP network that includes an HPNA LAN as the link between thehandset and the CM is provided. The clock in the cable modem is used tosynchronize transmissions of upstream packets to the DOCSIS MAC timing.Upstream transmission times are generally dictated by the DOCSIS headend equipment. In addition, for synchronous traffic flows, such as VoIP,the periodicity of the transmission of packets of the flow is directlyrelated to the upstream clock. Furthermore, the data samples in thepackets are acquired at a rate which is a derivative of the systemmaster clock. Because of these timing relationships, the cable modemclock must be synchronized to the clock in the cable modem head end. Atthe VoIP handset, the local clock is used to sample the analog voicechannel. This local clock must be related to the DOCSIS head end clockfor proper operation to occur.

As has been described briefly above, synchronization between clocks inVoIP handsets and CMs is necessary for two reasons: (1) the sample rateof the analog voice signal at the handset must match a standard 8 kHzvalue that is established for the entire voice transmission path inorder to avoid frame slips (lost samples or sample gaps) whichcompromise the quality of voice traffic and significantly reduce thethroughput of voice-band data flows; and (2) the framing of samples intoan RTP voice packet must occur synchronously to the arrival of anupstream grant at the DOCSIS MAC in order to minimize the latency of theupstream path. The SNR of the coded voice signal that traverses the PSTNmust meet the requirements of ITU-T recommendation G.712, whichspecifies an SNR of 35.5 dB for most input levels. Variation in the A/Dsample clock from a nominal 8 kHz frequency can be modeled as noise inthe coded signal, and therefore, a poorly tracking sample clock in thehandset can cause the handset to fall out of compliance with ITU-TG.712. The performance limits of G.712 translate directly into thejitter performance objective for the timing synchronization circuit ofthe HPNA VoIP system. A voice sample loss rate of 0.25 samples lost perminute must be maintained to support a toll-quality VoIP call. Thisrequirement translates into a long-term average tracking error of 0.52ppm between the handset and the CM. The overall latency that can beexperienced by a real-time interactive voice call before user-reporteddegradation of call quality occurs has been determined, throughexperimentation, to be no more than 150 msec according to ITU-Trecommendation G.114. Therefore, the one-way latency limit of 150 msecfrom ITU-T G.114 sets the performance goal for the latency requirementto be met by the HPNA VoIP system. The largest potential customer of thesystems to be built using the HPNA LAN for VoIP traffic has stated theirdesire for the final system to be capable of meeting the G.114 goal.

Both the CM and the handset will contain a local reference clock for theHPNA LAN. The two clocks must share a common value and must be runningat the same rate, averaged over time, with a maximum instantaneous errornot to exceed TBD, which matches the DOCSIS requirements. Severalmechanisms have been explored in order to solve the synchronizationproblem. Among them: (1) a software mechanism for determining thetimestamp at a remote location and correlating that time to the localtime, using round trip estimation to determine the correction forqueuing delay at each end, e.g., Network Time Protocol; (2) a relativeadjustment mechanism that sends only corrective indications between thetiming master and the timing slave. Both of these methods lack theability to discriminate between timing errors that are due to frequencydrift at the slave and errors that are due to inaccuracies indetermining the exact reference time. It is not well known if theinaccuracy of determining the reference time might create frequent andwide swings in the local reference clock, resulting in widely varyingsample intervals over relatively short periods of time, or worse,resulting in unstable clock behavior and frame slips. If wide or suddenvariations in reference time information is expected, then a reductionin tracking loop gain might solve the problem, but such a reductionmight place the tracking ability below the level where actual frequencydrift can be tracked well enough to meet the performance criteria forVoIP. However, the most compelling argument against a soft method oftime determination and tracking is the one that suggests that while thefrequencies in question may remain relatively stable over the periods ofinterest, the reference time establishment methodology (round trip timemeasurements) may not be very stable over short periods of time.Changing traffic patterns may produce sudden and persistent asymmetriesin the two legs of the round trip, resulting in a sudden change in thetimestamp estimation error. Without distinguishing the reference timeestimation error from the frequency drift error, it could be the casethat the DPLL inappropriately uses frequency corrections to adjust forthese sudden phase shifts. The sampling frequency could then be enoughout of step with the CM as to cause frame slips over relatively shortperiods of time. Voice-band data might suffer throughput degradationfrom the relative sampling time errors and voice traffic itself mightsuffer from harmonic distortions. The SNR requirements of ITU-T G.712might not be met. In any case, any of these methods ultimately requirethe implementation of a local clock generation circuit with a trackingfunction in order to create a clock source for the A/D circuit at thehandset. Given that the need for a tracking function is required, a moreformal mechanism for delivering precise reference time information isprovided in accordance with the present invention that does not confusefrequency drift with reference time estimation error.

The cable modems employ a DPLL to track the reference clock which islocated in the cable modem head end equipment. The performance of theDPLL must be sufficient to meet the requirements for digitized voicetransmission set forth in ITU-T recommendation G.712. ITU-Trecommendation G.712 gives an SNR of 35 dB to be maintained for PCMsignals. This value cannot be met with PCM μ-law encoding (beginningwith 12-bit linear samples) in the presence of more than about −70 dBnoise. The analysis done for the voice over DOCSI-S case, accounting forthe A/D and D/A performance, suggests that the output clock used forgenerating the 8 kHz A/D voice sampling clock should have a jitter of 5ns or less in order to meet these requirements. Any DPLL employed forclock tracking must be able to perform to this level if G.712 criteriaare to be met. Assuming that the highest sampled frequency in the voiceband is 4 kHz, then with 5 ns of jitter, a sine wave of 4 kHzexperiences a maximum instantaneous amplitude error of: 20*log [sin(5ns/250 μsec*2Π)−sin(0)]=−78 dB, a jitter of 30 ns produces an error of:20*log [sin(30 ns/250 μsec*2Π)−sin(0)]=−62 dB. The existing HPNA MACincludes a clock of 64 MHz, which could produce a jitter of 15.7ns:20*log [sin(15.7 ns/250 μsec*2Π)−sin(0)]=−68 dB. One further point tonote is that the CM device currently does not provide a straightforwardmeans for determining grant arrival times to the MIPS core. These factspoint favorably in the direction of at least a partial hardware solutionfor collection and delivery of grant and reference timing information.The general mechanism that is used to maintain timer synchronizationbetween the CM and the HPNA handset is very close to the method used bythe CM and the head end equipment in the DOCSIS network—however, as muchof the circuit as is possible can be implemented in software. Thisminimizes the impact to the MAC design while maintaining someflexibility in the design that allows the synchronization mechanism tobe fine-tuned outside of silicon development schedules.

As described above, the CM DOCSIS clock maintains synchronization withthe headend DOCSIS clock through the exchange of ranging messages andSYNC messages with the DOCSIS head end equipment. The timestamps inthese messages are inserted and extracted as the messages leave or enterthe DOCSIS MAC devices. The synchronization of the CM clock ismaintained by a circuit within the DOCSIS MAC called the TimingRegeneration Circuit (TRC). The CM extracts the timestamp from the SYNCmessage as the bits are arriving off of the wire. This timestamp ispassed to the TRC, where an immediate comparison to the local timestampis made. Any difference is used to adjust a DPLL which controls thelocal clock frequency. A ranging message is used to determine thetime-distance between the CM and the head end. The local clock isadjusted for this offset. The local clock in the CM is used to time CMDOCSIS operations, such as upstream transmissions. But CM VoIPoperations must also run synchronously to the DOCSIS head end clock, soa product implementation includes two functions which allow forPOTS/VoIP conversion devices (i.e. A/D and codec functions) to operatein synchronization with the DOCSIS clock. The first VoIP supportfunction of a product implementation is the export of a clock(TIC_CLK_OUT), which is a derivative of the local DOCSIS clock.TIC_CLK_OUT is used to drive the A/D sampling of the voice channel. Thisclock is used in order to insure that the sample rate of the A/D islocked in frequency to the DOCSIS clock. By doing this, the A/D samplingdoes not get ahead of or behind the DOCSIS grants—a situation whichwould result in lost samples or gaps in the stream of samples. Thesecond VoIP support function of a product implementation is the exportof a set of grant signals which indicate the arrival time of an upstreamgrant which corresponds to the desired framing interval of the collectedvoice samples. This grant signal indicates the framing boundary for aVoice over IP RTP data packet, which is a collection of A/D compressedand coded samples. An equivalent of these two functions is exported tothe HPNA LAN-attached handsets, in order to allow the analog portion ofthe handset to maintain a proper sample rate and to allow the DSP topacketize a set of samples in a timely manner, to avoid additional pathlatency.

The HPNA device does not need to duplicate the exact mechanism of theDOCSIS MAC device because the HPNA MAC at the CM has direct access tothe TICK-CLK-OUT clock. Therefore, a subset of the DOCSISsynchronization mechanism is implemented for the HPNA LAN MAC device. Inaddition, the HPNA LAN MAC mimics both the DOCSIS head end behavior andthe DOCSIS CPE behavior. The HPNA LAN MAC device located at the CM willprovide a timing reference to the HPNA LAN MAC devices located inhandsets. The CM's HPNA MAC will mimic the functionality of the head endequipment with respect to clock sourcing. That is, there will be amaster/slave relationship between HPNA MAC's in CMs and HPNA MACs inhandsets—the master dictates the current time to the slaves. Thisrelationship only slightly complicates the HPNA MAC time synchronizationsolution, as the same circuit can easily be made to operate in eithercapacity. The basic solution is similar to the DOCSIS MAC solution. ADPLL is incorporated within the HPNA MAC device. The DPLL is easilyprovided as a complete circuit (Timing Regeneration Circuit). Inaddition, the Smoothed TICK Clock Generator circuit is needed to producethe A/D sample clock at the handset side. In addition to the DPLL, theHPNA MAC includes a grant timing indication circuit. This circuit isbasically a timestamp function that operates whenever a grant issignaled by the CM. In practice, it is simply a modification to theexisting CM DPLL circuit. A few registers are added to the HPNA MAC tosupport the TRC operation, and a few more for supporting the GrantTiming Indication circuit. These registers are fully describedhereinbelow. The final modification to the HPNA MAC is to include up to6 new pins to provide an interface into the new circuits. In fact, thehandset requires only 2 pins to support the needed synchronizationfunction. The 6 pins is a maximum requirement for the timing masterconfiguration. The timing slave needs only 2 pins. A preferredembodiment is that the timing slave provide 3 pins. The pins employedfor the master functions do not need to be shared with the pins thatsupport the slave functions. The pins will operate differently dependingupon whether the MAC is at the CM or at the handset. The pins providethe functionality depicted in FIG. 78. There is some unsettleddiscussion surrounding the question of whether or not additional GrantPresent Indications are needed by the handset. That is, should thehandset HPNA MAC be capable of providing grant indications for more thanone VoIP connection? Because the current Broadcom CM reference designutilizes the MSI mode of the HPNA MAC device, the 6 pins can bemultiplexed with the upper AD pins of the PCI interface when in MSImode. It is not expected that other CM designs which might employ thePCI bus would also include the GrantRcv and reference clock signals usedby this interface. It is also not expected that PC-telephonyapplications need to be supported, therefore, the timing synchronizationfunction will not be available in PCI mode. One product requiring boththe use of the PCI mode and the grant synchronization interface has beensuggested. This product would be a PCI-based HPNA card for a PC, inwhich an RJ11 jack would be provided to allow for a single POTS lineconnection to the back of the PC. The card would serve a dual purpose ofproviding a data communications path for the PC while allowing the userto add a new VoIP line to his existing set of phone lines. This productimplementation would necessarily cost more than a stand-alone PCIdata-only card, since it would have to include the A/D, DSP, memory andmiscellaneous functions required to convert the POTS signal to HPNA. Inany case, if the reality of this type of product implementation isconsidered quite likely, then the PCI-based grant interface needs to befactored into the pin configuration of the PCI mode. In any case, if themost likely PCI-based grant interface scenarios represent only handsetapplications, then only three pins are needed to supply a completeenough interface. It may be possible to reduce this to two pins, if theDPLL input clock can be obtained from an existing, internal HPNA MACclock. At the CM side, the HPNA MAC uses the CM's TICK-CLK-OUT signal asthe reference input to the DPLL. Since this reference is already lockedto the head-end's DOCSIS clock, no corrections are ever needed for theDPLL that operates in the HPNA MAC at the CM site—it too runs in synchwith the DOCSIS clock. Note that no attempt is made to make the value ofthe CM HPNA MAC timer match the value of the DOCSIS MAC timer. This isnot necessary. However, it will be necessary to match the timer value inthe CM to the timer value in the handset. The synchronized referenceclock information needs to be transferred from the CM HPNA MAC to theHPNA handsets so that local sampling operations can maintainsynchronization with the DOCSIS reference, and so that the handsets canframe their samples to align with Upstream Grant arrivals.

The transfer of the CM HPNA MAC timestamp to the handset HPNA MAC timersis effected as follows. Instead of transferring DOCSIS SYNC-likemessages with timestamps inserted/extracted on the fly, the HPNAsynchronization mechanism relies on an internal MAC indication of framemovement to latch the current time into a timestamp register. The valuein the register is read and then delivered in a subsequent frame to thehandset which uses it to adjust its clock.

The CM HPNA MAC device is set up (through a register bit) to be a timingmaster, such that only transmit activity is timestamped. Ideally, onlyframes marked with the Timestamp transmit descriptor bit will cause theHPNA MAC timestamp to be latched. Software in the CM reads the timestampfollowing the sending of a frame that had the Timestamp descriptor bitset to TRUE. Software then constructs a TIMESTAMP REPORT messagecontaining the latched timestamp value and queues this frame for HPNALAN delivery to the broadcast address. The queue latency is unknown anddoesn't matter. The strict identity of the frame which generated thetimestamping event is unknown and doesn't matter, although it ispreferable to limit the frames which are timestamped. The mechanismchosen is to timestamp only TX frames that have the LTS descriptor bitset. To limit processing requirements at the receive end, the specialTimestamp Report Message (TRM) is defined. Only TRM will need to havetimestamp information recorded and delivered from the timing master.Timing slaves will then be able to ignore receive timestamp informationfrom all but TRM packets. Meanwhile, at the handset, and referring backto FIG. 80, the receiver is configured to act as a timing slave, suchthat only receive activity is timestamped. Each received frame triggersa timestamp to occur at the same relative position within a frame. Thereis a tradeoff wherein positioning the timestamp sample at an earlierlocation in the frame (up to and including the Type/Length field) yieldsa fixed offset from the beginning of the frame and results in theelimination of an offset correction. But the earlier timestamp allowsless time for the handset's logic to read the latched timestamp before anew frame possibly overwrites the latched value. A preferred methodcauses the latched timestamp to be incorporated within the RX statusword of each received frame, thereby eliminating any race condition. Inany case, the timestamp for each received frame is stored in memory.Associated with each timestamp is a TRM sequence number. The receivermay eliminate all RX status word timestamps that do not correspond toTRM packets. What remains is a database of TRM sequence numbers andtheir corresponding RX timestamps. When a TIMESTAMP REPORT messagearrives, the handset searches its local database for the referencedsequence number and compares the received timestamp with the storedtimestamp. The difference between the two values is used to determinethe DPLL error. The handset performs a filtering function on the error,adds the DPLL bias value and then writes the resulting value intoNCO_INC register 3014. In order to maximize the performance of the DPLL,it is recommended that TRM packets be sent in pairs. The rate oftransmission is suggested at about 1 pair per second. From the DPLL, anoutput can be fed to the pin output that will drive the codec of thehandset and ultimately, the A/D sampling circuit. Initialization of thehandset timer is achieved by accepting two TIMESTAMP REPORT messages,the second one of which refers to the first. The receiver adopts theerror indicated as an OFFSET value. This value is always added toreceived timestamps in order to calculate DPLL error. The DPLL counteris never modified. Since part of the DPLL loop is performed in software,the offset correction can easily be performed there. The CM HPNA clockis sampled as DOCSIS upstream grants arrive. The grant arrival times isthen communicated to individual handsets through HPNA packets, in orderto allow the assembly and queuing of RTP voice packets to be scheduledto insure that the packets will arrive at the CM just in time for thenext upstream grant. Packet assembly overhead, queuing latency,transmission time, and CM packet processing time is subtracted from thegrant time in order to generate a packet assembly start time thatinsures that the packet meets the next upstream grant at the CM. Themechanics of this operation are as follows. DOCSIS upstream grants aresignaled by the cable modem through the GrantRcv[4:0] interface.GrantRcv[4] is used to indicate the arrival of a grant from the headend. GrantRcv[3:0] are used to signal the SID which corresponds to thecurrent grant. Each SID corresponds to a particular connection flow,such as an individual call flow. The timing of the arrival of each grantneeds to be communicated to the appropriate handset. In order toaccomplish this, the 5 GrantRcv signals are fed to the CM HPNA MAC, andthe HPNA MAC's internal timestamp value is latched whenever theGrantRcv[4] signal becomes active, provided that the GrantRcv[3:0]signals match the value set up in the tscSID register of the HPNA MAC.The MIPS core of the CM programs the tscSID register to match the SIDcorresponding to the call in progress for a given handset. Once theGrantRcv[4] timing is latched in the HPNA MAC, the MIPS core reads thelatched timestamp and subtracts worst case queuing latency, transmissiontime, and CM packet processing time. It then sends a GRANT_TIMESTAMPmessage to the appropriate handset. A SID to MAC address mapping existsat the CM in order to allow for proper grant timing signaling. This mapis constructed and maintained by the MIPS core. The handset receives theGRANT_TIMESTAMP message (an extended version of the TIMESTAMP REPORTmessage). The handset adds N*T time units (N=integer, T=RTP packetperiod) minus packet assembly processing latency to the timestamp fromthe message in order to calculate a time that is in the future. It thenloads this time into the GRANT_TIME register so that the HPNA MAC canproduce a grant-sync output to the codec at the appropriate time. Whenthe TRC reaches GRANT_TIME, the GrantRcv[4] signal is asserted for oneclock pulse duration and GRANT_TIME register 3030 is automaticallyincremented by the value in GRANT_PERIOD register 3017. A register bitexists to disable the generation of grant pulses on GrantRcv[4]. Asafety bit is used to indicate that the grant time has been indicated,in order to prevent the case of a grant time having been passed beforeit was programmed, and hence, no grant signals ever being generated. Thesafety bit would be a register bit that changes from a 0 to a 1 when thegrant time is signaled on the output pin, and which can only be reset to0 by software. Note that the timing master must switch between transmitand grant-arrival timestamp latching operations. The implementation mayinclude either one latch that is switchable between the two functions,or two latches to satisfy both requirements. The receive frame timestamplatching operation may share one of the latches mentioned, or it may beseparate.

Referring to FIG. 79 the PINS associated with HPNA MAC changes aredepicted. The device is either a timing master or a timing slave, butnever both. Therefore, the maximum number of pins required for eithermode is 6. This requirement is for the timing master, where the MSI modeis expected to be employed. Newly defined registers for the HPNA MAC areprovideed. These registers do not come with the TRC circuit.

-   NCO_INC[15:0] written with the filtered difference between slave and    master time plus NCO bias value when tracking adjustments are being    made to the DPLL-   tscSID[3:0] determines which Grant[4] input pulses will cause a    timestamp latch event—latch events only occur when Grant[3:0] match    tscSID[3:0] AND Grant[4] is asserted AND tMastertMaster is TRUE AND    sGrant is true-   GRANT_TIME[15:0] contains a time that is to be matched against the    slave time+offset_adjust. When a match occurs, Grant[4] output is    asserted for one clock pulse and the value of GRANT_TIME is    automatically incremented by the value of GRANT_PERIOD (multiple    registers to support multiple channels?)-   GRANT_PERIOD[15:0] (fixed at 10 msec, so not needed?)-   TX_TIMESTAMP[31:0] contains timestamp latched as a result of a    transmit event (e.g. preamble transmitted AND TIMESTAMP bit of TX    descriptor is TRUE?) (shared with GRANT TIMESTAMP register)-   RX_TIMESTAMP[31:0] contains timestamp latched as a result of a    receive event (e.g. DA=BCAST?), the lower 16 bits of this value will    be automatically stored in the RX status word-   V_SCALE[7:0] scaling value to be applied to the timestamp clock in    order to produce the required A/D voice sampling clock-   TS_SCALE[7:0] scaling value to be applied to the NCO output clock in    order to create a common Timestamp clock frequency    Miscellaneous register bits could go into existing registers if    needed.-   EN_REF_OUT when set, this bit enables the V_CLK_OUT and Grant[4:3]    output drive functions. This control bit only causes these pins to    become outputs when the chip mode is MSI.-   S_EXT_REF_CLK when set, the TRC circuit input reference clock source    is the DPLL_REF_CLK pin, when reset, the TRC input clock source is    internal to the device-   tMastertMaster used to switch between latching timestamp on transmit    signal instead of receive signal, default value is    tMaster5Master=TRUE-   sGrant used to switch between latching timestamp on Grant[4] signal    instead of on transmit signal-   GRANT_SIGNALED needed to make sure that the Frame[0] signal was    actually asserted—the slave controller may have set a GRANT_TIME    that was not sufficiently far in the future, due to processing    latency—if the GRANT_TIME value had already been passed when it was    loaded, then no grant signals are being generated externally—this    bit can be used to verify that the GRANT_TIME value has been reached    (is this necessary?—our only timing problem would be the cycles    between receiving the GRANT_TIMESTAMP message and calculating a    future time, then loading the GRANT_TIME register . . . no queuing    latency is involved). This bit is resettable by the host.-   S_DPLL_OUT when set, this bit causes the V_CLK_OUT mux to use the    DPLL output clock directly, without passing through the two integer    dividers.-   S_NCO_TS Used to select the NCO output, or the second integer    divider output as the clock which drives the Timestamp counter. When    this bit and S_REF_TS are both set to 1, then the NCO output clock    is used to drive the timestamp counter. When this bit is set to zero    and the S_REF_TS bit is set to one, then the second divider output    is used to drive the timestamp counter. Default value is ONE.-   S_REF_TS Used to select between the NCO reference clock input, or    the output side of the NCO as the clock which drives the timestamp    counter. When set to 1, selects the NCO reference clock input as the    source clock for the timestamp counter. The timestamp counter must    have a reference clock input of 4.096 MHz. Default value is ZERO.-   NCO_RESET When set to one, this bit causes the NCO counter to be    reset to x00000000. The NCO is not normally reset, even during a    hard reset of the chip. The lack of a natural reset for the NCO is    to insure that there is always a clock output at V_CLK_OUT. The use    of the NCO_RESET bit should be restricted to test environments,    since it is likely to cause a glitch on the V_CLK_OUT signal. Note    that NCO_RESET MUST NOT BE TIED TO PIN RESET, since this would    prevent V)CLK_OUT from running during a board reset.    TX Descriptor bits include:-   LTS Latch TimeStamp: causes a timestamp latch event on transmit    frames when this bit is set to 1.    RX Descriptor bits include:-   RXTS[31:0] 32-bit receive timestamp value

FIG. 80, which, depicts components of an embodiment an HPNA TRC circuitin accordance with the present invention, is now described in moredetail. Adder 3010, reference clock signal 3012 and NCO 3014 areprovided. An output from the NCO 3014 is fed into integer divider 3016.This clock in the slave device gets divided down to 8 kHz (V_CLK_OUT)3018 since it is running at much higher speed to maintain an accuracy.The V_CLK_OUT feeds the sampling circuitry of the CODEC. The softwaremakes a determination as to whether the clock is running fast or slowvia SNOOP_BUS 3020 which is located inside the transceiver which allowsthe software to communicate with the hardware. The PCI bus writes avalue to register 3022. Synchronizer 3024 is provided to make sure thatthe change in register 3022 is synchronous to the NCO 3014. The outputclock gets speeded up or slowed down depending on the value loaded intoregister 3022. The software looks at the timestamps that are received atthe slave and determines if the slave clock is running slow or fast. Itmakes an adjustment to the register 3022 value which adjusts the speedof the NCO 3014. It does this typically every one second, or whatevertime is necessary for a defined accuracy.

There are two other aspects on the receive side for the slave. When thepacket comes in, every packet creates a signal which samples the currentvalue of the timestamp clock which is running based upon the DPLL. Thesampled clock is put into a structure that is associated with thereceive packets. Every received packet has clock timestamp associatedwith it. The software that has the responsibility of identifying thespecial packets that contain timing information from the master and fromthose it can look to see what the time it received those packets was andit can see what time the master sent them. The master will have sent apacket that will, when it gets sent, get a timestamp associated with it.The software goes and reads the timestamp and puts it into a follow-uppacket. The protocol involves the sending of two packets. The firstpacket from the master gets a timestamp stored locally and the packetgoes out without having a timestamp included in it. It makes it acrossto the converter. The converter takes a timestamp on the same packet.Both the master and slave have taken a timestamp. However, neither knowswhat the other's timestamp is. The master then reads the timestamp outof the register and puts it into a follow-up packet and sends it along.The follow-up packet doesn't get timestamped by anybody. The follow-uppacket arrives at the slave device. The slave device now has the timethat the first packet was sent out and the time that the first packetwas received. Once that information the slave can then see thedifference between them.

The grant timing that is determined from the DOCSIS network is delivereddirectly to the transceiver for the HPNA. That information is gatheredby the timestamp circuit on the master and input to the circuit viaGrant (4)timing signal, with S_GRANT enabling the path. Grant [3:0]allows multiple different grant identifiers (one of sixteen) to beselected. When the interested in grant identifier sees it's grant, thatlatches the timestamp. Therefore, when a grant occurs there is atimestamp associated with the grant at the master. The master then readsthat timestamp information, puts it into a packet and delivers thatpacket with the grant timestamp identifiers associated with it to allthe nodes. The node associated with that particular grant identifierpicks up the information and now it knows when its grant occurred. Itwill have been able to relate its time to the master's time by lookingat the offset between the time it received according to its clock andthe master's time. For example, using human time differentials, if themaster indicates that it sent a packet at 12:00 o'clock, and the slaveindicates that it received the packet at 3:30 o'clock, it knows that thetwo clocks differ by 3½ hours. Since it knows that it is 3½ hours off,then when the master latches a grant time in its timestamp register,when it delivers that time the slave then knows that it needs to adjustthe time by 3½ hours to make it updated to its local time. Once it knowsthe local time of the grant, then it adjusts that backwards by the timeit needs to assemble the packet and deliver it on the HPNA network. Itworks backwards to figure out what the latest time is that it shouldsend that packet out of the network. It puts that time into a GRANT_TIMEregister 3030 and when the local time in the slave matches at aexclusive-OR comparator 3032 an output signal Frame[0] is created whichgoes to the voice CODEC and tells it to deliver 80 samples. In fact, thesignal Frame[0] can be sent to any portion of the circuit which ismaking the actual decision as to when to call a set of 80 samples aframe. The circuit also automatically updates the grant time periodrate, e.g., 10 msec, such that when the grant time matches the currentlocal time, 10 msec is automatically added to the grant time and 10 mseclater another match of the grant time with the current local time andthe framing signal will be created again.

Of note is that the NCO error input is calculated by the device driver.The BIAS is added to the error, and the driver writes the resultingvalue to the NCO_INC register 3022. The correct BIAS value depends uponthe V_CLK_OUT frequency requirement for the specific application. TheV_CLK_OUT signal must be square (50% duty cycle). The V_CLK_OUT signalwill begin with a default rate at power up. During RESET, the rate willbe fixed. After RESET, the software will write values to various controlbits that may change the rate of the V_CLK_OUT signal. These changesmust not produce glitches on the V_CLK_OUT output. The circuit asdepicted allows V_CLK_OUT frequencies in the range: mear DC to 100 MHz.However, because of the requirement for the timestamp to be running at4.096 MHz, an additional requirement must be placed on the V_CLK_OUTsignal. The V_CLK_OUT signal must either be a ratio of integers divideof 4.096 MHz, or it must be a ratio of integers multiple of 4.096 MHz,where the integers must be in the range of 1–255, inclusive. This shouldprovide sufficient range of V_CLK_OUT operation for all expectedapplications. The accuracy of the DPLL decreases as the output frequencyis reduced because the rounding error remains constant in magnitude,while the control word value decreases in magnitude. For a directconversion of 200 MHz to 8 kHz, the control word for a 32-bit DPLL is29F16, which produces a rounding error of 4 ppm. If this rounding erroris unacceptable, then any of several remediation steps can be taken,including, adding bits to the DPLL register. Adding 2 bits to theregister changes the error to 1.1 ppm. Another option is to perform lessconversion in the DPLL, then feed the DPLL output to a divider to getthe final output. It turns out that additional divide steps are requiredanyway, because a fixed rate clock is required for the timestampfunction. The fixed rate for the timestamp is chosen to be 32.768 MHz.(If the timestamps at the master and slave differ by a power of two,this would be acceptable, since software could accommodate thedifference. Some other integer relationships are easy to adapt in asimple CPU—for example, the factor of 6 is easily obtained by twoadditions.) The chart set forth in FIG. 81 shows the jitter in the DPLLoutput when the reference clock is 200 MHz and the DPLL output clock(CNT[31]) is 32.768 MHz. The jitter variance is +/−2.5 ns and thefrequency of the jitter is about 3.3 MHz. The jitter frequency is wellabove the audio range, and the +/−2.5 ns causes noise that is below −70dB in amplitude, thereby allowing the A/D to achieve the required 35 dBSNR requirement of ITU-T recommendation G.712. Lower frequencycomponents do exist in the jitter waveform, but the amplitude of thesecomponents is significantly lower than the 3.3 MHz signal. The offset ofthe jitter shown in FIG. 81 is corrected over time by DPLL frequencyadjustments, such that the offset will ultimately vary around 0.

Referring back again to FIG. 80, to determine the master timestamp,DPLL_REF_CLK 3040 is a fixed clock provided for register 3014. It isconsidered the “master clock” to which other devices are to besynchronized. After dividers 3016 and 3019 divide the signal fromregister 3012 to provide TS_CLK 3042 which drives timestamp register3011, which is the source of the timestamp for the packet. The output oftimestamp register 3011 is provided to TX_TSTAMP register 3044 whichtakes the timestamp in response to its EN becoming active. EN becomesactive when TX_SIG 3046 is asserted at a fixed point in thetransmission, e.g., at the end of preamble. The output of TX_TSTAMPregister 3044 is made available to software through register access onthe device.

Still referring to FIG. 80, on the slave side receives a packet. Thetimestamp at reception has no known relationship to the master timestampother than counting at the same rate. Analogous to the timestampoperation described above, when RX_SIG 3048 is asserted at a fixed pointin the transmission, e.g., at the end of preamble, which agrees with themaster side fixed point, which enables the load operation of RX_TSTAMP3013 of whatever is then in its TIMESTAMP register 3011. The output ofRX_TSTAMP register 3013 is similarly made available to software.

Referring back to the master aspect of FIG. 80, the transmitter softwarereads the latched output of TX_TSTAMP 3044 and puts the value into asubsequent packet and sends the packet along to the slave device. Theslave device receives the sent packet it reads its latched output fromRX_TSTAMP 3013 and determines when the event occurred on the receiveside.

Referring again to FIG. 71, the relationship is shown between when apacket is told that there is an opportunity to transmit on theasynchronous network and when the packet is created and queued onto thetransmitter's asynchronous MAC device. Time point 3050 depicts thetransmit opportunity on the asynchronous network. It involves adding upthe sum total of delays encountered by the creation of the packet at theslave device on the asynchronous network and its delivery to the DOCSISMAC and transmission on the DOCSIS network. The timespan between timepoint 3050 and timepoint 3052 is the delay that occurs within the cablemmodem's queue. The timespan between timepoints 3052 and 3054 is the timeit takes for the cable modem to process the frame. The timespan betweentimepoints 3058 and 3056 is the propagation delay for the HPNA MAC. Thetimespan between time points 3058 and 3060 includes queuing delay andany wait because of transmission activity on the wire. Given thetimespans the slave device creates the packet to align with the transmitopportunity.

In a preferred embodiment, LAN delivery latency is improved byconverting the typical collision resolution algorithm from a randomassignment to a fixed backoff, as in accordance with the presentinvention. The collision resolution algorithm provides a random number(0, 1, 2) after having a collision on the network. The random number isused to build a tree of all the colliding devices until there isestablished one branch of the tree that has only one device on it and heis then free to transmit without experiencing a collision. By having thetiming master communicate tree branch information to each of the devicesthat wish to participate in synchronous-timing, and assign the randomnumbers to choose when there is a collision, ahead of time, the masterin effect has established a tree resolution with the minimum number ofcollisions possible.

Referring again to FIG. 80, NCO incrementer 3022, in response to errorinput from software, filtered an biased by software, adjusts the feed ofthe count of NCO 3014. This helps compensate for drifting frequencybetween slave and master. With NCO incrementer 3022 set to the nominalreset value, NCO 3014 halves the frequency of the DPLL reference clock.TS_SCALE register 3070 and V_SCALE register 3072 along with integerdividers 3016, 3019 are used to allow at the slave side differentcrystal frequencies that don't match the crystal frequencies at themaster side. The outputs from NCO 3014 and dividers 3016 and 3019provides clock 3018 which feeds the CODEC clock which takes samples ofthe analog stream, the dividers helping create a slower clock for theCODEC. Further, signal Frame [0] signal 3074 is also provided to theCODEC to indicate to the CODEC when to slice off a set of samples forpacketization, based upon the transmit opportunity times as to when aset of samples is to be assembled into a packet. GRANT_PRD register 3017is loaded with signals representative of the periods of the transmitopportunities. When GRANT_TIME register 3030 initial grant time loadedbecomes the same as TIMESTAMP register 3011, a true compare output isprovided to enable a reload of GRANT_TIME register 3030 to reload granttime plus grant period output from 32 bit adder 3076. With thecomputation of the grant period offset, the next transmit opportunitytime in the future for a transmission to occur is provided, and signalsthe CODEC that a time has arrived to assemble a packet for queuing fortransmission.

Still referring to FIG. 80, with regard to the master side operation SGRANT signal 3078 is an enabling signal and Grant [4] 3080 is receivedfrom the DOCSIS side of the network, a synchronous timing event. Whenthis occurs the current timestamp is latched into TX_TSTAMP register3044.

Referring to FIG. 82 a limited HPNA TRC embodiment is shown. Thisimplementation will allow a timing master to be fully implemented. Atiming slave will require an external DPLL and external grant signalinglogic or a software approximation of grant signaling. (A softwareapproximation of grant signaling would mean that software sets a timerto be interrupted when the next grant time arrives. The timer is setbased on a read of the current timestamp as compared against theexpected next grant time. The software would either initiate the framingand queuing process upon interrupt, or it would generate an outputsignal through a general purpose pin to cause external logic to createthe frame. The accuracy of the grant timing on the slave device is notas critical as that required for maintaining a proper sample rate, sincethe queuing and contention delays are very highly variable anyway.) Thetiming slave will have a single input, which is the DPLL_REF_CLK. In theembodiment, the timing slave output pins are deleted. In a timing slaveconfiguration, the DPLL can be external to the device. The PINS andvarious Bit Locations are depicted in FIGS. 83 a–83 g. A new TXDescriptor bit for the embodiment includes:

Bit 25 LTS LatchTimeStamp: causes a timestamp latch event on transmitframes when this bit is set to 1 New RX Descriptor bits for theembodiment include: Byte 27 rxTimeStamp[31:24] MSbyte of rxTimeStampByte 26 rxTimeStamp[23:16] upper middle byte of rxTimeStamp Byte 25rxTimeStamp[15:8] lower middle byte of rxTimeStamp Byte 24rxTimeStamp[7:0] LSbyte of rxTimeStamp

The circuit embodiments in accordance with the present invention requiresoftware control to complete the timing synchronization function. Withthe same circuit, HPNA network nodes are able to operate as one of twotypes at any given time. Nodes will either function as a timing master,or as a timing slave. There may be more than one timing master active atany given time on a particular HPNA LAN. Timing master and timing slavenodes have different physical connections and are serviced by softwarein differing manners. The behavior of the software algorithm for eachtype of node is described hereinbelow.

The timing master will perform the following tasks:

-   1. Initialize the device as a timing master-   2. generate pairs of TRM packets at 1 second intervals-   3. generate a pair of TRM in response to a received TQM-   4. generate a TRM in response to the establishment of a new channel    for a given MAC address, or in response to a received TSM (TRM in    this case does not need to be a pair)-   5. generate a TRM with the lost-lock indication when lock has been    lost at the Cable Modem or other source of reference timing    information (such as a DSL modem)    To initialize the timing master, the tMaster bit of the control    register is set to force the device to operate as a timing master.    The sGrant bit of the control register is reset. TRM sequence number    space t x0000 is initialized. TRM pairs are sent using a period of    at most one second. TRM pair generation is as follows. A TRM message    is created with TRM_type=x00 and with TRMSeqNum set to the next    unused TRMSeqNum. PrevTRMSeqNum is set to x0000. Timestamp is set to    x00000000. NumGrants is set to x00. Destination address is fixed as    the broadcast address. The TRM is queued in the TX queue of the    embodiment with the LTS descriptor bit set to 1. After the TRM is    reported to have been transmitted, the value latched in the    TX_TIMESTAMP register is read. A new TRM with TRM_type=x00 is    created with TRMSeqNum set to the-next unused value. PrevTRMSeqNum    is set to the value of TRMSeqNum in the first TRM of the pair.    Timestamp is written with the value of TX_TIMESTAMP that was just    read from the embodiment. NumGrants is set to x00. DFPQ priority of    all TRM is set to 6. The second TRM in the TX queue of the    embodiment is queued with the LTS descriptor bit set to 0. The    reception of a TQM is a request by a timing slave for the immediate    transmission of a pair of TRM. The master responds by immediately    executing the TRM pair generation procedure. The normal 1 second    periodic timer is not disturbed. A TRM may include Grant Timing    information. Not all TRM are required to include grant timing    information. A TRM with grant timing information is generated in    response to either of two events: (1) a latency-sensitive service    flow is initialized (e.g. a VoIP connection is established); or (2)a    TSM is received. In either case, the TRM is constructed in the    following manner. First, Grant timing information is obtained. The    timing master keeps a list of MAC addresses and their associated    SIDs. SIDs are Service Flow ID's that are assigned by the cable    modem head end equipment when the VoIP connection is set up. The    cable modem software must track all currently active SID values and    keep a table which associates each value with an HPNA LAN MAC    address. When a TSM is received, the timing master must get all    channel ID's associated with that MAC address and then gather grant    timing information for each channel ID. Grant Timing information is    obtained through the following mechanism. The driver insures that no    outstanding LTS bit remains set in the active TX descriptor list. A    selected channel ID (SID value) is placed into the tscSID register    of the embodiment. The current value of the TX_TIMESTAMP register is    read and stored. The sGrant register bit is set. The driver waits 10    msec (or whatever time is appropriate for the given channel—the wait    time is equal to the period of the traffic flow). The driver reads    the TX_TIMESTAMP register and compares it to the stored value. If    the-values differ, then the driver assumes that a valid timestamp    has been captured for the selected SID. If the values are the same,    then the driver waits for the period of the flow and reads the    TX_TIMESTAMP again. The sGrant register bit is cleared. The TRM is    constructed as follows. A TRM message with TRM_type=x00 and with    TRMSeqNum set to the next unused TRMSeqNum is created. PrevTRMSeqNum    is set to x0000. Timestamp is set to x00000000. NumGrants is set to    x01. Destination address is set to the broadcast address. MACAddr is    set to the MAC address of the requesting node. Channel_ID is set to    the appropriate channel ID. Gtimestamp is set to the value read from    the TX_TIMESTAMP register. The LTS bit of the TX descriptor is set    to 0. DFPQ priority of all TRM is set to 6. The driver may choose to    collect grant timing information for multiple channel_ID's for a    given MACAddr before creating a TRM with grant timing information.    However, it is best to deliver the grant timing information for any    channel as quickly as possible. Note that the tscSID register is    loaded with a different value depending upon whether the device is    attached to a BCM3308 or a BCM3350 cable modem device. BCM3308 SID    values are positionally coded in the tscSID register, e.g., SID    value of x3 corresponds to tscSID value of x8. SID values are    directly represented in the tscSID register, e.g., SID value of x3    corresponds to tscSID value of x3. There needs to be an indication    from the master reference clock source indicating a loss of lock.    When this occurs, the master follows the same procedure as for    sending TRM pairs, but with the TRM_type set to x01 instead of x00.

Timing slave devices will receive clock and grant timing informationfrom timing master devices. Timing slaves will use this information fortwo purposes. The clock information will be used to keep the local clocklocked to the master clock. The grant timing information will be used todetermine when to frame a set of voice samples and send the frame to theCM.

There are several local variables to be maintained by the slavesoftware. They include: NCO_BIAS—the nominal divider for the NCO thattranslates the 200 MHz reference crystal to the timestamp clockfrequency (nominally 32.768 MHz); SLAVE_OFFSET—the difference betweenthe master clock timestamp value and the slave timestamp value;Frequency_adjustment—the long-term estimate of the slave's frequencyerror from the master reference smoothed with a filtering function;integrator_gain—coefficient for smoothing of the frequency_adjustmentterm; Phase_adjustment—the instantaneous adjustment to the slave'sfrequency error from the master reference, multiplied by the linear_gainterm linear_gain—coefficient for smoothing of the phase_adjustment term.The detailed relationships of these terms will be explained hereinbelow.The timing slave is initialized as follows. The tMaster bit of thecontrol register is reset to force the device to operate as a timingslave. The NCO_BIAS is set to the value of

${NCO\_ BIAS} = \frac{2^{32}*f_{TS}}{200}$where f_(Ts) is equal to the desired Timestamp frequency in Megahertz.f_(Ts) is fixed at 32.7668 for this application. With this value forf_(Ts), the NCO BIAS is a x29F16B12. The frequency_adjustment is set toZERO. The integrator_gain term is set to 0.02 (TBD xxxx). Thephase_adjustment is set to ZERO. The linear_gain term is set to 0.90(TBD xxxx). The SLAVE_OFFSET is set to ZERO. With regard toinitialization of frequency_adjustment, in order to allow for frequencysynchronization, the timing slave device incorporates a DPLL. The DPLLreference input has a nominal frequency of 200 MHz. The reference clockdrives an NCO which yields a clock with a reduced frequency which isintended to track the master's clock. The initial BIAS value for the NCOwas calculated based on the assumption that the reference clock is atexactly 200 MHz and the master clock is running at exactly 32.768 MHz.However, the actual reference clock value is only nominally equal to 200MHz. The typical crystal supplying the slave reference time has an errorof +/−100 ppm. This error offset is measured, and the NCO-BIAS valuemust then be corrected for this error. The local reference frequencyerror can be measured directly by simply comparing the master's TRMinterval measurement with the slave's. When any TRM pair arrives, themaster will indicate the current time. With knowledge of the master timefrom a previously-received TRM pair, it is possible for the slave todetermine the amount of time that has passed, assuming that the master'sclock is correct. Then the slave can examine its own estimate of thetime that has passed during that same interval to determine the localerror. If M_(x) is the master timestamp at time T_(x), and S_(x) is theslave's timestamp value at time T_(x), then the following equationdescribes this method:

${{Slave\_ Frequency}{\_ Error}} = {\frac{S_{2} - S_{1}}{M_{2} - M_{1}} - 1}$Since the error could be quite small, the slave will have to wait for along enough period of time to accurately measure it. With the timestampaccuracy at 30.5 ns (at each end, using 32.768 MHz as the timestampclock), each reported timestamp can be inaccurate between 0 and 0.06usec. Assuming a required tracking error of less than 1 ppm, the slavewould have to measure the master/slave time difference over an intervalgreater than 0.06 μsec/1 ppm=0.06 seconds=60 milliseconds in order toinsure that the frequency error had been measured to greater than 1 partin 100. I.e. after 60 msec, the frequency drift error contribution wouldbe 6 usec and the measurement error would be −0/+0.06 usec. It isconvenient to wait much longer than this, so that the error contributiondue to timestamp resolution is greatly reduced. If the slave waits thenormal 1 second TRM interval, then the measurement error is very smallcompared to the maximum desired tracking error of 0.52 ppm. (Themeasurement error falls to than 0.06 ppm.) In any case, the first stepfor the timing slave is to wait for the arrival of two pairs of TRM.When the first pair of TRM arrives, the timing slave stores the materand slave indicated timestamps and waits. (The first TRM of the pairyields a slave timestamp, the second of the pair reveals the mastertimestamp for the same event.) When the next pair of TRM arrives, theslave calculates the slave frequency error as described above. Adivision operation is necessary for the calculation, but the divisiononly needs to be performed during initialization. The operation is nottime-critical. The frequency error needs to be translated to an NCO BIASadjustment value in order to allow the NCO to be adjusted to the properfrequency. The result is the initial value for the frequency-adjustmentvariable: Frequency_adjustment=NCO_BIAS*Slave_frequency_error. Theintegrated_gain term is not applied during the initialization step. Thefrequency_adjustment will be added to the NCO_BIAS term and thephase_adjustment term to create the NCO control word. An additionalerror exists because the master timing reference has some non-zeromeandering component which is due to the cable modem's attempts tomaintain frequency lock to the head end timestamps. Once the cablemodem's clock is locked, this meandering should not exceed about 1 ppm.The error is small enough to ignore during the initialization step—afterinitialization, we can assume that the slave and master are closelylocked. The remaining error will disappear in a short time during thetracking phase. Timestamp acquisition is the process whereby the timingslave determines the relative offset between the local time and themaster time. Timestamp acquisition at the timing slave node is performedas follows. Once the frequency_adjustment has been initialized, themaster and slave timestamp clocks are declared to be in sync. Therefore,the indicated master and slave timestamps for the second received pairof TRMs that was used to calculate the initial frequency_adjustmentvalue give the nominal clock offset. This offset is stored in theSLAVE_OFFSET variable and is used by the slave to calculate any neededreference times. SLAVE_OFFSET=S₂—M₂. The SLAVE-OFFSET value is not usedto modify the DPLL, nor is it used to modify the slave's timestampregister. SLAVE_OFFSET will never be updated, because the DPLL willattempt to track the master timestamp and keep the offset constant. Anymaster time that is signaled to the VoIP circuit (such as a grantindication to determine framing) will be converted to an equivalentslave time first by adding the SLAVE_OFFSET value, and then the slavetime will be signaled to the VoIP circuit. Note that under normalcircumstances, the timing slave will return a timestamp for every RXframe. The timing slave preserves the timestamp which corresponds to themost recently received TRM frame in order to be able to calculateinterval durations as needed. The initial phase_adjustment that would becalculated from the second pair of TRM would be zero, because the masterand slave are declared to be locked in phase at that point in time (i.e.at initial sync time). As a result, there is no phase_adjustmentnecessary until the third pair of TRM is received—and even then only ifa measurable error has accumulated. So the initial value of thephase_adjustment term remains ZERO. The initial NCO control word iscalculated with the initial frequency_adjustment and phase_adjustmentterms along with the NCO_BIAS value:NCO_Control=NCO_BIAS+frequency_adjustment+phase adjustment. TheNCO_control word is written to the NCO control register at thecompletion of the initialization step. In the BCM4220, the NCO is notimplemented. The NCO control register is external to the device. Thetracking function measures the error from the most recent TRM intervaland then attempts to correct for that error in the next TRM interval.The error is corrected by modifying the frequency and phase adjustmentterms based on the current error and then updating the NCO control word.Following the arrival of any TRM pair, the current slave timestamp erroris determined: Curr_slave_error=S_(x)−M_(x)-SLAVE_OFFSET. Where S_(x) isthe slave timestamp for the current TRM pair and M_(x) is the mastertimestamp for the current TRM pair. For each TRM interval, the intervalduration is determined: Curr_interval=M_(x)-M_(x-1). The phaseadjustment for a given interval is calculated as follows:Phase_adjustment=linear_gain*NCO_BIAS*curr_slave_error/curr_in terval.The frequency adjustment for an interval is calculated as follows:Frequency_adjustment=frequency_adjustment+int_gain*NCO_BIAS*curr_slave_error/curr_interval, where int_gain=integrator_gain.One could continue to use the equation:

${{Slave\_ Frequency}{\_ Error}} = {\frac{S_{x} - S_{x - 1}}{M_{x} - M_{x - 1}} - 1}$to determine the frequency error for a given interval and thensubstitute this value for the curr_slave_error/curr_interval term in thegiven frequency_adjustment equation. But thecurr_slave_error/curr_interval term gives an adequate approximation,even with aggressive values for the integrator_gain term. The assumptionis that the slave remains fairly well-locked to the master, and in thatcase, the approximation holds. By using only one equation, an extradivide operation is avoided. After modifying the adjustment values, theNCO control words is recomputed and reloaded into the DPLL:NCO_CONTROL=NCO_BIAS+frequency_adjustment+phase_adjustment If the timingmaster creates TRM intervals of consistent 1 second times (with lowjitter), then an additional math operation can be avoided by assumingthat the curr_interval value is always equal to 1 second. Given that theTRM frames are sent with LL priority 7 (=DFPQ priority 6), the deliverylatency jitter of a TRM should be well below 10 msec with 99%confidence. If a TRM pair is missing, then the original math operationneeds to return, since the next interval will be an integer multiple of1 second, requiring division by something other than 1. (As a furthersimplification, errors measured during longer intervals could beignored, thereby avoiding this problem.) There is the possibility ofmissing timestamp messages during normal tracking. The separation ofcrystal offset error from master-slave drift, NCO rounding error andreference source jitter is required in order to allow for free-wheelingNCO operation when no correction information exists for an interval.During intervals for which a TRM pair is lost, the NCO is clocked at thenominal NCO BIAS plus the frequency error adjustment, i.e.,phase_adjustment is reset to ZERO. The frequency adjustment isunmodified in such circumstances. When a valid pair of TRM does arrive,the phase error that accumulated during the free-wheeling operation willbe corrected in roughly a single TRM interval (depending upon thelinear_gain term). The chart depicted in FIG. 84 a shows the performanceof the circuit with the following parameters:The timestamp clock frequency is 24.576 MHz.The nominal TRM interval is 1.0 sec.The linear gain is 0.9 over the nominal TRM interval.The integrated gain is 0.1 over the nominal TRM interval.The number of TRM pairs that arrive at the slave correctly is 95%.The jitter in the master clock is +/−1 ppm corresponding to +/−1 sigma,using normal distribution.TRM interval jitter is corrected in making phase and frequencyadjustments.The simulation models a master clock jitter which is probably worse thanwill be encountered in reality, since the master clock will be createdby a DPLL with correction intervals of 200 msec (MAX), while thesimulation assumes master clock corrections which occur at 1 secintervals. In the real system, the higher correction rate for the masterclock will likely cause smoothing of the master clock jitter as observedby the slave. Also, it is expected that the CM clock will contain muchless than 1 ppm jitter over intervals of several seconds. In general,the behavior of the circuit is very good, with the jitter shownfundamentally reflecting the jitter in the master clock input signal,with some amplification due to the timestamping inaccuracy and the factthat the slave system can only correct for past errors. It is impossibleto construct a circuit which anticipates and corrects for future masterclock jitter. Note that in all cases, the behavior of the circuitmodeled is to not offer a phase correction in the absence of anyreceived TRM.

The chart in FIG. 84 b shows the tracking behavior of the DPLL whenthere is no master jitter, as a means of illustrating the performance ofthe DPLL in the presence of a stable master reference. Note the twoorders of magnitude change in the vertical scale from FIG. 84 a. In thecase when the cable modem completely loses lock, communication from thecable modem to the head end is disallowed. When this loss ofsynchronization occurs at the timing master, lost-lock TRMs will be sentto timing slaves so that they do not attempt to track the master clock.When the timing master re-acquires lock, the master must resume sendingTRMs with a locked indication. Timing slave devices noting thetransition from lost-lock to locked state must perform a new acquisitioncycle. During the period of lost lock, the slave may choose to continueto send the VoIP frames, since the master may recover quickly enough tosend some of them. With regard to the reception of grant timestamps, theGRANT_SIGNALED bit is firt cleared to zero. The timing slave adjusts thereceived grant timestamp value with the SLAVE_OFFSET value. An integermultiple of the grant period is added to the result and the final valueis written to the GRANT_TIME register. The software sets timer for justover one grant period. After the expiration of the timer, the softwarechecks the GRANT_SIGNALED bit. If set, then the grant is being properlysignaled to the framing logic. If not set, then the software must addadditional integer multiples of grant period to the originally receivedgrant timestamp value and repeat the previous steps. In the embodimentof the present invention, the grant signaling logic is absent. In thiscase, the grant timing is approximated by a software timer which isbased on the estimated time to the next grant. The grant indication(framing) output would be signaled through a general purpose I/O pin.

The Timestamp Report Message protocol is intended to convey system-leveltiming information between two nodes of an HPNA network. One node isassumed to be the timing master, and the other node is a timing slave.There may be more than one timing slave for a given timing master.Timing master devices send timestamp messages to timing slaves on aperiodic basis. Timing slaves use the timestamps to synchronize a localclock to the timing master's clock. The TRM protocol also supports theconveyance of specific time information relating to connection-basedservice flows. In particular, the desired arrival time of a packettransferred from timing slave to timing master may be conveyed to atiming slave device through the TRM protocol.

The TIMESTAMP REPORT message (TRM) is a Link Control Frame ofSStype=TBD, is set forth in FIG. 85(1)–(3). A pair of timestamp reportmessages (TRM) is sent every 1 second to allow for timing recovery. Whenthe first message of each pair is sent a timestamp is recorded as themessage is being transmitted onto the medium by the timing master. Theexact moment at which the timestamp for the TRM is sampled is notimportant—however, the consistency of the sample time is important. AllTRM timestamps are taken at a fixed time (master_timestamp_offset)relative to the time at which the first preamble symbol is transmittedonto the wire. The variation in the value of master_timestamp_offset canbe no more than +/−2 μsec. The absolute value of master_timestamp_offsetmust be greater than or equal to ZEROμsec and less than or equal to 64μsec. The timestamp that was recorded during the transmission of thefirst TRM of a pair is placed into the body of the second TRM. Thesecond TRM is transmitted as soon as is possible following the firsttransmission. The second TRM of the pair does not require a timestamp tobe recorded. The number of Slot Timestamps in a TRM may be zero. It isassumed that Slot Timestamp periods for each channel have beencommunicated through an out of band mechanism. All timestamp protocolmessages are sent with link layer priority of 7, which corresponds toDFPQ priority of 6 for all possible mappings. Timing slave devicesnoting a transition of master state from lost-lock to locked initiate anacquisition cycle when the transition is noted.

The TIMESTAMP Request message (TQM) is set forth in FIG. 86. Thetimestamp request message is sent by a timing slave to request thedelivery of a pair of TRM. TQM messages are always sent to the broadcastDA, since only one timing master is active on any HPNA LAN segment.

The TIMESTAMP Slot Request message (TSM) is set forth in FIG. 87. Thetimestamp slot request message is sent by a timing slave to request thedelivery of a set of TRM which contains a lot timestamp for each of theactive channels associated with the requestor's MACAddr. The set of TRMthat is sent by the timing master-in-response to the receipt of a TSMmay consist of a single TRM, or it may consist of more than one TRM. TSMmessages are always sent to the broadcast DA, since only one timingmaster should be active on any HPNA LAN segment.

VoIP HOL Blocking Solution Implementation Requirements

As discussed above, a backing away from the randomization process ofcollision resolution is needed in order to provide the best possiblequality of voice service. In essence, the present invention provides amechanism for selecting and distributing a pre-defined ordering ofcollision resolution rather than using a randomly derived ordering. Whenit is done in this manner, in essence a dictated deterministicresolution is layed on top of the distributed network. This is done justin the context of voice. Other type of traffic does not have an issuewith resolution as it currently stands. This portion describes thegeneral requirements of the operation of the HOL blocking within a VoIPsystem in accordance with the present invention. VoIP frames are notspecifically identified to the device driver—however, all VoIP framesare identified by a higher layer and assigned the LL priority of 6,which translates to DFPQ PRI=7 for all possible mapping combinations.Therefore, all VoIP frame queuing rules are generalized to include allDFPQ PRI=7 frames. DFPQ PRI=7 frames have priority access to thenetwork. DFPQ PRI=7 frames have priority access over all TX queues thatlie in the path to the network. This includes any and all TX queues thatlie within the device driver. If a case exists where multiple driverqueues contain DFPQ PRI=7 frames, these frames are passed to the MACdevice in the order that they are received in the aggregate. HW-basedLICF frame generation is not enabled when DFPQ PRI=7 frames may be inthe TX queue, or may be expected to arrive for queuing from higher layersoftware. The FLUSH command will not remove HW-generated DFPQ PRI=6 LICFframes from the TX queue in the hardware. Because HW generated LICFframes are not flushed, they will continue to block higher-priorityframes. Note however, that the FLUSH command WILL REMOVEsoftware-generated DFPQ PRI=6 LICF frames from the hardware TX queue.Once a connection is established, the arrival of DFPQ PRI=7 frames willbe periodic. There may be times when no DFPQ PRI=7 frame exists in anyqueue (i.e. software queues and hardware queues). If the softwareexpects that additional DFPQ PRI=7 frames may be arriving within thenext 1 second, then HW-based LICF generation must not be enabled. Theeasiest test would keep HW LICF generation off unless the device driverhas determined that the system should be steeped. DFPQ PRI=7 frames arere-ordered. This is a general rule that applies to all traffic (with theexception of LARQ retransmissions). This rule continues to be valid forVoIP traffic. DFPQ PRI=7 frames include a LARQ header. DFPQ PRI=7 framesare transmitted twice as per a control switch. The second frame appearseffectively as a LARQ-induced retransmission, even though NO NACK wasreceived to prompt it, i.e., the second transmission has the samesequence number as the first transmission, but the LARQ_RTX bit is setto 1. However, the second transmission is placed into the outgoing TXqueue at the same time as the first frame is placed into the queue—i.e.there is no delay between the queuing of the first copy and the secondcopy of the frame. For ordering purposes relative to other DFPQ PRI=7frames, the original and the copy is treated as an inseparable pair. Acontrol switch is present which allows this function to be enabled ordisabled. Any received LARQ NACK frame referencing a previouslytransmitted DFPQ PRI=7 frame is ignored as per a control switch. Framesthat have been flushed (blocking frames) are re-queued if the returnedstatus indicates that the flush was effective. Frames which are notflushed are not re-queued. Software makes the determination of thedisposition of all frames in the queue according to the returned statusof each frame. Any frames that have been determined to have been flushedare re-queued. When re-queuing flushed frames, original queue orderingare preserved within a given level of DFPQ priority. Re-ordering offrames of differing priorities is allowed and encouraged. The ITU G.712specification for total distortion is shown in FIG. 88 which includesthe error introduced by the non-linear quantizer (Compander). It wasfound, using a Matlab simulation of the Compander and an ideal uniformquantizer, the SNR associated with the compander and the uniformquantizer at full scale and at −30 dB input level are 38.5 dB and 36 dB,respectively. The compander SNR is roughly independent of input signalstrength from full scale to −45 dB, because the quantization noise poweris proportional to the signal strength. Therefore, the SNR of the ADCand DAC must be high enough to avoid dominating the compander noise.FIG. 88 indicates the total SNR must be greater or equal to 35 dB withinput range from 0 dB to −30 dB. FIG. 89 a shows the required SNR of theADC/DAC based on our Matlab simulations in order to meet the ITU G.712specification. Although the required ADC and DAC SNR is less than 60 dBwith input full scale. Nevertheless, it can be seen that the ADC/DACmust have at least 14 bits of resolution since a noise floor of −84 dBis required. This is consistent with the ITU's choice of a 14 bit inputrange for the compander. Our simulation showed that for a −40 dBm inputwith 13 bits of resolution the SNR was 25.4 dB, the required SNR is 29dB. For a 14 bits of quantization, the resulting SNR is 30 dB. If ajitter clock is used for the sampling of the input signal, then it isnecessary that the SNR associated with the jitter clock is much lessthan 60 dB. The effect of the jitter clock can be considered as asinusoidal time jitter with amplitude K and frequency W. This will causea sampling of the input signal at T+K*sin(WT) instead of at time %. Ifthe input signal is a sinusoidal signal A*cos(woT), then the effect ofthe jitter clock is the same as the input were A*cos {W0(T+K*sin(WT)}.

Also, A*cos {Wo(T+K*sin(WT)}=A*cos {(WoT)+T*K*sin(WT))};

If T*K<<1, then the jitter will cause the output to have sidebands atWo+W and at Wo−W with an amplitude of A*K*W0/2.

Thus, the SNR is A*K*Wo/A=K*Wo which is normalized to the input signallevel A. For K=5 ns and Wo=2*pi*4 kHz, then K*Wo=1.2566e 4 (−78.0158 dB)is the noise level relative to any input level. FIGS. 89 b and 89 c showthe total SNR associated with the sinusoidal and random noise models ofthe jitter clock, respectively. It can be seen that the increase in thetotal SNR for either the sinusoidal or white noise jitter models is lessthan 0.15 dB.

The effect of the jitter clock (24 MHz) based on its characteristics andthe way it is used in an analog test chip is now described. This jitterclock is derived from a NCO inside the DOCSIS MAC and transceiver. Itscharacteristics are plotted in FIG. 90. The fundamental frequency ofthis waveform is about 200 kHz which is based on 1000 cycles of theinput 200 MHz clock to the NCO. The effect of the jitter can beconsidered as a 200 kHz sawtooth time jitter with amplitude +/−2.5ns.Again, the jitter output of a sinusoidal input A*cos(2*pi*Wo) can bewritten in the form of A*cos {WoT+Wo*F(T)}, where F(t) is the sawtoothwaveform. For K*WO<<1, the jitter will provide sinusoidal signals atfrequency near 200 kHz and their harmonic components. FIGS. 91 and 92show respectively the ADC and DAC data paths of the analog test chip. Inthe ADC data path, the jitter clock can be modeled with the 200 kHzsawtooth signal as input to the digital CIC decimator. Similarly, in theDAC data path, the jitter clock also can be modeled with the sawtoothsignal at the output of the noise-shaping digital modulator. The CICdecimator will eliminate all signal with frequency above 48 kHz. Notethat the input to the CIC decimator and the output of the digitalmodulator are noise-shaping signals. A simulation in Matlab with a 3 kHzsinusoidal input, showed that any signal outside the 48 kHz frequencyband will be considered as the out-of-band noise. Therefore, the jittersawtooth at 200 kHz in the ADC path will be completely removed by theCIC decimator. For the same reason, the jitter sawtooth in the DAC pathwill be completely filtered by the analog lowpass filter.

Collision Signal Slot Assignment

The delay associated with transmitting a frame on an HPNA V2 network hasthree major components: serialization delay (the time it takes foractual transmission of the frame header and data), deferral delay(s) dueto waiting for frames to be transmitted (at any priority), and collisionoverhead when multiple stations wish to send at the same priority (whichmay include one or more deferrals). In order to provide high qualityVoice over IP (VoIP) service, it is necessary to control the maximumlatency of frames containing voice sample data. Frames normally contain10 ms worth of data, and are therefore sent once every 10 ms. Per ATTVoIP requirements, the overall end-to-end delay for voice sample datafrom microphone to speaker must be 150 ms or less. One prospectiveallocation of the end-to-end delay budget for a system providing voiceover cable service provides (at most) 10 ms of delay beyond thecustomer-premises cable modems, which results in at most 5 ms for thetransmission of voice data over a local network such as an HPNA V2network. If VoIP frames are the only frames sent at the highest prioritylevel (7), then a given VoIP frame need wait for at most one lowerpriority frame to complete plus the time it takes to resolve collisionsbetween multiple VoIP stations for the right to transmit. The delaywaiting for a prior transmission to complete can be up to 3.2 ms (thetime to send a maximum size frame at the minimum rate of 4 mbps),although field trials show that most home support payload rates of 10mbps, with a corresponding maximum frame transmission time of 1.2 ms.However, with a default HPNA V2 implementation it is not possible toestablish a strict upper bound on the time it takes to send a frame,even at the highest priority level. The default implementation of thecollision resolution mechanism includes a “random” function for theselection of one of three signaling slots used to establish precedencefor the next transmission among the colliding stations. If two or morestations choose the same signal slot following any collision, thenanother collision will occur followed by another collision resolutioncycle. The result of this mechanism is that there is no upper bound onthe maximum number of collisions that can occur before all competingnodes in the original collision complete a successful transmission. Asolution to this problem is relatively simple. Using the protocoldefined below, stations that intend to generate low-latency VoIP (andsimilar) traffic are pre-assigned their signal slot selections for up toeight rounds of collision resolution, to be used only for priority 7frames. Careful assignment of these values guarantees a minimum upperbound for the number of collisions: one for two or three stations, twofor up to five stations, three for up to seven stations and four for upto nine stations. (Note that only two rounds of signal slot values areneeded for up to nine stations).

A set of values for Collision Signaling Slots is called a CSS sequence.The set of CSS sequences can be enumerated, and each sequence has anexplicit “rank” in an ordered tree structure that determines the orderof frame transmission when a collision occurs with one or more stationsthat also have [unique] CSS sequence assignments. Although the basicfunction is simply assigning CSS sequences to stations that send VoIPtraffic, the protocol needs to address a number of goals: (1) Providespecial handling for multi-channel stations. Up to three multi-channelstations should be assigned CSS sequences that differ in the first slotid, so that when traffic for additional channels is sent following thefirst round of collisions, additional collisions (due-to two or moremulti-channel stations) will be minimized. (2) The protocol shouldhandle optimized CSS sequence assignments for stations with activechannels. It may, or may not, be advantageous to assign CSS sequences toall VoIP-capable nodes. However, since the total number of VoIP stationsmay far exceed the number of stations with active channels (the designgoal for home operation is four active, full-duplex, VoIP channels),dynamic assignment and/or reassignment is highly desirable. (3) Theprotocol should allow for operation in the absence of a centralizedsequence assignment authority (i.e. a CSS master node). In thisenvironment, individual client nodes are allowed to assign their own CSSSequence values, advertise them, and reassign them if necessary, inorder to avoid using identical sequences.

In addition to the CSS Protocol itself, one new bit-flag(CSS_Master_Capability) is defined for advertisement via the CSAprotocol. The flag advertises that a station can operate as a CSS masternode. It does not indicate that the advertising node is necessarily thecurrent CSS Master. It only serves to indicate capability. Note alsothat VoIP nodes implement the CSA protocol in order to dynamicallyadvertise the use of Link Layer (LL) priority 6, that corresponds tovery low latency (<10 ms) traffic and is mapped to physical priority 7for transmission over the phoneline. CSS client nodes utilize thepresence or absence of the CSA Master Node flag in the current set ofreceived flags to determine the method by which CSS sequences areassigned. When no master node is present, clients broadcast a requestfor the current CSS sequences of other clients (sent as replies) andthen send an announcement choosing an unused CSS sequence. With a masternode on the network, clients request CSS sequence assignments and waitfor the response of a master node.

In addition to the CSS_Master_Capability flag, the CSA message ismodified by including a TLV (Type/Length/Value) extension to theexisting fixed fields. The TLV-extension is used to request andacknowledge the exchange of CSS Sequence values among nodes.

Now turning to collision signaling slot assignment protocol, a CSSSequence is eight two-bit values concatenated: two-bit values in therange [0,2] indicate a specific signaling slot, to be used following acollision, while a value of 3 indicates the use of a randomly selectedvalue chosen by the client at the time of the collision. If a nodeencounters 9 collisions, selection reverts to a random algorithm untilthe frame is either transmitted or dropped. A CSS Master (also CSSCurrent Master) is a node which accepts the responsibility forassignment of CSS sequence values to CSS clients. Some nodes may haveCSS master capability yet may not be operating as the CSS master at anygiven point in time. Only one CSS master is allowed control of thenetwork at a time. There may be transitional periods of overlap betweenmultiple masters. A CSS Client is a node which may request theassignment of a CSS sequence. A CSS client may choose its own CSSsequence in the absence of a CSS master. A CSS master may also operateas a client. In such a case, the request for a CSS Sequence is not sentto other nodes, but the acknowledgment of the CSS sequence is sent toother nodes.

With regard to CSA Extension to support CSS Frame types, theCSA_Master_Capability flag is added to the CSA message's flag set, inthe Flags 1 octet. The flags set forth in FIG. 93(1)–(2) are used forCSA_CurrentTxSet, CSA_OldestTxSet, and CSA_CurrentRxSet in Capabilitiesand Status Announcement control frames. All CSS (Collision SignalingSlot) protocol messages take the form of a CSA message (see HPNA V2characteristics) with one or more CSA extension TLVs included. A CSAextension TLV is a Type/Length/Value field which follows the fixedfields of the CSA message. The CSA Extension follows the last fixedfield of the CSA frame (CSA_CurrentRxSet), and precedes thenextEthertype field of the CSA frame. The CSA Extension for CSS has theform set forth in FIG. 94. A CSS frame is built from the CSA using theCSS CSA Extension subtype. All CSS messages are sent as CSA messageswith at least one embedded CSA Extension of subtype CSS. A CSS requestmessage is a CSA message with: The CSS_Master_Capability flat set toeither ZERO or ONE; (CSS Master capable nodes which are not operating asthe current CSS master must act as CSS client nodes, but they still settheir CSS_Master_Capability flag.); At least one CSS TLV with thefollowing values: CSEType=x00, CSELength=x08, CSS_MAC=MAC address of therequesting client, CSS_SEQ=xYYYY, where YYYY has a value in the rangexFF01–xFFFE, and where the least significant 8 bits correspond to thenumber of active link layer priority 6 channels sourced by therequesting client. An active channel is one for which some non-zero flowof traffic is currently being generated. CSS requests are sent by a CSSclient whenever the number of active link layer priority 6 channelschanges. A CSS assignment message is a CSA message with:

The CSS_Master_Capability flag set to ONE; (Only the CSS master can makeassignments. In the absence of a master, individual nodes can only makerequests, which should be respected by other CSS client nodes, but mightnot be.) At least one CSS TLV with the following values: CSEType=x00,CSELength=x08, CSS_MAC=MAC address of the client to which the sequenceapplies,CSS_SEQ=xYYYY, where YYYY has a value in the range x0000–xBFFF.The CSS Assignment may contain multiple CSS TLVs, indicating multipleassignments. In addition, the CSS assignment always contains a CSS TLVwith the CSS sequence for the CSS master, if one has been assigned.(i.e., the CSS Master's assignment messages always contains the CSSmaster's own CSS acknowledgments.).A CSS acknowledgment message is a CSAmessage with: The CSS_Master_Capability flag set to either ZERO or ONEas appropriate—both CSS clients and CSS masters may send CSSacknowledgments; At least one CSS TLV with the following values:CSEType=x00, CSELength=x08, CSS_MAC=MAC address of the client to whichthe sequence applies, CSS_SEQ=XYYYY, where YYYY has the value of theCSS_SEQ as assigned to the client by the CSS master, and where YYYY hasthe value “FFFF” when the acknowledgment is in response to a CSSS dropmessage. The CSS acknowledgment is always sent by a CSS client inimmediate response to the reception of a CSS assignment to itself, andthenceforth in all CSA messages that are normally generated by the CSAprotocol. The CSS acknowledgment with CSS_SEQ=XFFFF is always sent by aCSS master in immediate response to the reception of a CSS drop messagefrom a client. In such cases, the CSS_MAC value carries the MAC addressof the CSS client that sent the CSS drop message. If a CSS master nolonger requires a CSS sequence, it sends a drop acknowledgmentreferencing its own MAC address. This is done to keep the CSS sequenceinformation in synch at other potential CSS master nodes. A CSS dropmessage is a CSA message with: The CSS_Master_Capability flag set toZERO or ONE; (Master capable nodes acting as CSS clients may send CSSdrop messages. The current CSS master never sends a CSS drop message.)At least one CSS TLV with the following values: CSEType=x00,CSELength=x08 CSS_MAC=MAC address of the client to which the sequenceapplies CSS_SEQ=xFF00. The CSS drop message is sent by a client which isterminating all active link layer priority 6 flows and no longerrequires the possession of a CSS sequence.

Master nodes respond to received client request messages by sending anassignment message. Master nodes will have the complete list of activeCSS Sequences and therefore will not err by assigning the same sequenceto more than one requesting client node. Master nodes may reassign thesequence for any node in an unsolicited manner for purposes of grantingan earlier-resolving sequence to multiple-channel nodes, or for otherpurposes (e.g., collapsing the outstanding sequence tree as active nodesbecome inactive). Master nodes age the received active node CSSinformation at the same frequency as other CSA information. Master nodesalways send their own sequence value (should they possess one) inoutgoing CSA messages, just as clients do. This announcement serves thepurpose of informing other potential masters, of all sequencesoutstanding. This information is useful, should a potential master needto replace the current one. When a master receives a CSS drop messagefrom a client, the master responds by sending a CSS acknowledgmentmessage containing a CSA_SEQ of xFFFF for the dropping client.Similarly, when the master node deletes a client from the assignedsequences list due to aging, the master node sends a CSA messagecontaining a CSA_SEQ of xFFFF for the dropped client, to indicate thatthe client has been dropped. Again, this unilateral indication servesthe purpose of keeping all potential master nodes' assigned sequenceinformation coherent. It also allows the CSS client the opportunity tore-request, should the unilateral drop acknowledgment have been made inerror. In the special case where the current CSS master drops its ownrequirement for a CSS sequence, no CSS drop message is sent, but anacknowledgment of the drop is sent by the master in order to informother nodes of the change in the outstanding sequences, i.e., the CSSmaster sends an acknowledgment for its own drop, but the drop message isnot sent. The master for any system is determined by mastershipcapability indication in the CSA flag set, and by the MAC address ofeach potential master. The node indicating mastership capability withthe lowest MAC address is always the selected master. If a node appearsin a system, and the new node has mastership capability, then the newnode does NOT advertise its mastership capability and it does NOTperform master functionality until a full CSA aging period has elapsed.Note that CSA messages should still be sent, but theCSS_Mastership_Capability flag must not be set. This insures that thenew master does not inadvertently gain the current master position untilit has acquired all relevant CSS sequence information which may alreadybe present in the system. However, there is an allowed acceleration ofassertion of the mastership capability flag. This occurs in the casewhen the new master can determine that there is no current master in thesystem. A new potential master node can quickly make this determinationthrough any of several means including: The new potential master nodesends a drop message and does not receive a drop acknowledgment. (Thistest is repeated several times to be certain that either the drop or theacknowledgment has not been simply lost). The new node observes the lackof master acknowledgment to other clients' request/drop traffic. The newnode sends a CSA request and notes the lack of any received mastershipcapability indicating in all received CSA messages. This test isrepeated several times to be certain that either the request or theresponses have not been simply lost. In any case, if the new potentialmater node can reasonably assume that no master is currently present inthe system, then it may cancel the normal waiting period and immediatelyadvertise CSS master capability and immediately assume the role of theCSS master. It is possible in such a situation, that the client nodes inthe system may have assigned their own sequences in the absence of amaster. When the new master asserts itself, it attempts to collect theset of self-assigned sequences before making its own assignments. Thenew master may unilaterally re-assign sequences to each client. Once anew potential master with a lower MAC address has collected a completeset of CSS sequence information, or a new potential master hasdetermined that no current master exists, it announces its mastercapability by setting the CSS master capability flag in all subsequentoutgoing CSA messages. The existing master (if any) will recognize thepresence of the new master and relinquish mastership, but continue toadvertise its own CSS mastership capability, after verifying the factthat the new master's MAC address is lower than its own. There may becases where the previous master fails to immediately recognize the newmaster, and in such cases, a client may receive multiple CSS Sequenceassignments. The client replaces its existing sequence with the newestsequence and immediately generates a CSA acknowledgment of the CSSSequence. The new master repeats its CSA advertisements as often as itdeems necessary in order to get the previous master to finally recognizeit as the new current master. It is possible that a potential masterhas, through missed CSA frames, aged the current master's information,and has assumed the current master position even though it has a higherMAC address. That is, the new master believes that the rightful masterhas quietly exited the network. If this occurs, then the current master(if still present) must defend its mastership by specifically sendingCSA messages at an unspecified higher rate, and by correcting any clientassignments that the incorrect master may have made. The usurping masterwill see both the re-assignments and the repeated master CSA messagesand back down. If the current master disappears, then all potential newmasters will recognize which has the next lowest MAC address and allwill defer to that node. If the current master doesn't respond torequests and/or drops from clients, then all potential replacementmasters may prematurely age the current master and the next master inthe line of succession assert its right to mastership and beginresponding to the clients.

Client nodes request a sequence from the master node by sending a CSAcontaining a CSA Extension of subtype CSS to the broadcast address (CSSrequest). The client node places its own MAC address into the CSS MACfield and sets the CSS_SEQ value equal to xFFyy, where “yy” correspondsto the number of channels actively transmitting link layer priority 6frames. Client nodes acknowledge receipt of the CSS master's sequencevalue by repeating the assigned sequence value in the CSS_SEQ field ofall subsequent outgoing CSA messages. (Note that all subsequent CSAmessages will contain a CSS CSA Extension.) The CSS_MAC field is set tothe client's MAC address. The repetition of the sequence owned by eachclient serves to prevent the aging of the client's information at themaster node. It also allows a potential replacement master to havecomplete sequence assignment information in case it is called upon toreplace the existing master. If the number of active link layer priority6 channels for a client node changes, then the client node sends a newCSS request message to indicate the change. The CSS master may or maynot modify the client's CSS sequence. In either case, the CSS masterresponds with a CSS assignment in order to acknowledge the receipt ofthe CSS request. When a client node discontinues sending all traffic atlink layer priority 6, then it sends a CSA frame containing a CSS CSAExtension subtype, with the CSS_SEQ value set to xFF00 and the CSS_MACvalue set to its own MAC address (a CSS Drop message). Effectively, theclient is advertising for zero channels of traffic at link layerpriority 6. The client continues to advertise this value for CSS_SEQuntil the master acknowledges receipt of the frame (through the CSS_SEQvalue of xFFFF), or until no master is present in the system, asdetermined by CSA aging at the client. In the case when no CSS master ispresent (as confirmed by the lack of a received CSS_Master_Capabilityindication in all received CSA messages), the client node claims a CSSsequence by choosing a sequence and sending a CSS acknowledgmentmessage. All subsequent CSA messages contain the same CSSacknowledgment. If one client node chooses the same sequence as anotherclient node, then the new claimant to the sequence has priority for thatsequence. A specific algorithm for choosing a sequence in the absence ofa CSS master is not specified, but such an algorithm includes factorssuch as: outstanding sequences in use and number of channels active byeach CSS client. The original owner of the sequence must choose a newsequence. A good example algorithm for choosing sequences is as follows.All client nodes monitor all CSS exchanges and keep a list of in-usesequences. Normal CSA information aging supplies to CSS information.Client nodes are divided into two general classes: single channel andmulti-channel link layer priority 6 sources. Multi-channel sources areafforded relatively higher positions in the ordered tree created by theset of sequences of collision resolution. If a client node requires aCSS sequence and is a single channel source, then it chooses the nextunused sequence in the ordered tree. If a client node requires a CSSsequence and it is a multi-channel source, then it chooses the nextsequence following the last sequence used by the list of multi-channeldevices. This choice is made, even if it conflicts with an existingsingle-channel device. If a multi-channel device drops itself (or isaged) from the set of used sequences, then the lowest-orderedmulti-channel device claims the abandoned sequence. If no multi-channeldevice exists at a lower point in the ordered tree, then the lowestordered single-channel device fills the abandoned space. If asingle-channel device drops itself (or is aged) from the set of usedsequences, then the lowest ordered single-channel device claims theabandoned sequence, unless the abandoned sequence is lower in theordered tree. It is recommended that self-assigned sequence values donot exceed 4 levels in depth (i.e. CSA_SEQ should have the form xYYFF,where YY has any hex value). In any case, any colliding sequences amongclient nodes will be afforded a new opportunity to resolve randomlyafter all 8 signal slot values have been used, since the node behaviorfollowing the use of all 8 2-bit values is to revert to random selectionuntil either the frame is successfully transmitted, or the frame isdropped at the transmitter. While a client does send a drop asappropriate in the no-master case, without a master, there is noacknowledgment for the drop, and therefore, the drop is repeated severaltimes in order to insure reception by other clients. However, in theevent of the failure of any of the clients to recognize the explicitdrop, the drop will be recognized in time through the aging process.

The MAC hardware supports the CSS protocol by providing a 16-bitregister (CSS register) which is loaded with the CSS_SEQ value from theCSS message. Whenever the frame at the head of the transmit queue is alink layer priority 6 frame (highest priority on the physical network),the 16-bit register becomes the source for signal slot selectionfollowing link layer priority 6 collision events in which this node wasan active transmitter involved in the collision.

In the unmodified version, the signal slot value is always chosen atrandom. For the HPNA V2 implementation, a random number in the range[0,2] is used. The selected number is used to determine during whichsignal slot the colliding node signals to indicate its participation inthis round of collision resolution. With the CSS assignment scheme,succeeding 2-bit values from the CSS register are used in place ofrandom selections. In this way, collision resolution will be ordered,rather than random. This allows an upper bound to be placed on theresolution of any collision. The value of the upper bound is a functionof the number of nodes participating in the collision and the specificCSS values that each participating node possesses. Because each 2-bitvalue can represent 4 possible integer values, and because the HPNA V2protocol requires selection of an integer signal slot value from a rangeof only 3 values, the 4th value is used to revert to random selection ofthe signal slot number (for the given collision). The table set forth inFIG. 95 indicates the desired codings for the CSS register bits. Aninitial collision for a frame causes the 2 most significant bits of theCSS register to be used as the signal slot integer selection for thatcollision. Successive collisions encountered by transmission attemptsfor the same frame use successively lesser significant 2-bit values fromthe CSS register. If a frame encounters 8 collisions, then all possiblenon-overlapping 2-bit values will have been used, and the signal slotinteger is chosen by random selection. Whenever a new link layerpriority 6 frame arrives at the head of the transmit queue, then thesignal slot selection returns to the most significant 2 bits of the CSSregister, regardless of how far through the CSS register a previous linklayer priority 6 frame's signal slot selection may have proceeded.

The PSD mask specified is such that compliant transmitters should beable to meet FCC Part 68 Section 308-e-1-ii.

The mask also specifies a limit of −145 dBm/Hz below 2.0 MHz, whichensures compatiblity with G.992.1, G.992.2 and ISDN.

The mask includes a notch covering the Radio Amateur bands between 7.0and 7.3 MHz which reduces the maximum PSD to −85 dBm/Hz. This is lowerthan the VDSL recommendations for PSD in the amateur bands. Since theVDSL spectral compatibility has been developed over the last severalyears in several standards bodies, including the ITU, this spectral maskshould be compatible with RFI emission requirements in countries outsideNorth America, such as UK, Japan, Germany and France.

Mode Selection

An HPNA V2 device is capable of acting as an HPNA V1 transmitter andreceiver when required by other devices on a network. The HPNA V2transceiver complies with the document “Home Phoneline NetworkingAlliance HPNA V1 PHY Specification V1.1” when trasmitting and receivingHPNA V1 frames, with the following additional guidelines:

-   -   1. An HPNA V2 device is configured by default as low-power and        high-speed, per the HPNA V1 specification.    -   2. The implementation of HPNA V1 high-power mode in an HPNA V2        device is not required, will not be certified, and is not        encouraged.    -   3. The use of high-power mode, if implemented in an HPNA V2        device, is not recommended for remediation of HPNA V1 network        problems.

The HPNA V2 compatibility mode pulse must not be used when transmittinga true HPNA V1 frame.

When operating on a network that has mixed HPNA V1 and HPNA V2 stations,an HPNA V2 station uses Compatibility Mode. In this mode, HPNA V2stations use the media access control algorithm defined in “HomePhoneline Networking Alliance HPNA V1 PHY Specification V1.1.”

The format of frames transmitted by an HPNA V2 station varies in HPNA V1Mode, Compatibility Mode, and HPNA V2 Mode as follows:

-   -   1) An HPNA V2 station in HPNA V1 Mode transmits only HPNA V1        format frames with PCOM=1 or 2.    -   2) An HPNA V2 station in Compatibility Mode,        -   a) transmits HPNA V1 format frames to broadcast, multicast,            HPNA V1 stations, stations of unknown type, or HPNA V2            stations under conditions specified in Section 2.3.3.1 of            “Interface Specification for ILine10 HPNA V2 Technology Link            Layer Protocols,” (the Link Layer Specification). The PCOM            shall have the value 1 or 2.        -   b) shall transmit HPNA V2 Compatibility format frames to            HPNA V2 stations as permitted by Section 2.3.3.1 of the Link            Layer Specification.    -   3) An HPNA V2 station in HPNA V2 Mode shall transmit only HPNA        V2 Native format frames.

All HPNA V2 stations are able at any time to identify and receive framesin any of the following format: (a) HPNA V1 format frames, (b) HPNA V2Compatibility format frames, (c) HPNA V2 Native format frames.

When stations transmit HPNA V1 format frames, they shall code the PCOMfield as specified in FIG. 96

All HPNA V2 stations shall power up in HPNA V2 Mode. In order todetermine HPNA V1 Mode or Compatibility Mode, HPNA V2 stations shallkeep the internal boolean state variables specified in FIG. 97. Therelative precedence of the variables in mode determination is alsospecified in FIG. 97, (a) being the highest, and (4) the lowest.

While in HPNA V2 Mode with Link Integrity Status=DOWN, an HPNA V2station that detects an HPNA V1 format frame with PCOM Station Type=1asserts V1_DETECTED. V1_DETECTED is cleared if a 2 second period elapseswithout detection of any frames with PCOM Station Type=0.

An HPNA V2 station that detects an HPNA V1 format frame with PCOMStation Type=0 (see FIG. 4.1) asserts V1_DETECTED. V1_DETECTED iscleared if a 60 second period elapses without detection of anysubsequent HPNA V1 format frames with PCOM Station Type=0.

An HPNA V2 station that detects or transmits an HPNA V1 format framewith PCOM Station TYpe=2 asserts V1_SIGNALED. V1_SIGNALED is cleared ifa 60 second period elapses without detection or transmission of anysubsequent HPNA V1 format frames with PCOM Station Type=2.

Each HPNA V2 station combines the Capabilities and Status Announcement(CSA) information received from other stations using the logical orfunction, to set internal state variables ConfigV1, ConfigV1M2, andConfigV2.

An HPNA V2 station determines HPNA V1 Mode or Compatibility Mode withthe following logic, implementing the precedence specified in FIG. 97:

-   V1M2_MODE:=(not ConfigV1) and ((not ConfigV2) or ConfigV1M2) and    -   (ConfigV1M2 or V1_DETECTED or V1_SIGNALED)-   1M8_MODE:=ConfigV1-   10M8_MODE:=not (Compatibility_MODE or HPNA V1_MODE)

Future specifications can use the Frame Type (FT) and Payload Encoding(PE) fields to define new frame formats and new modulation types/rates.The etiquette for sharing the 4.5–9.5 MHz channel are defined by thevalid Carrier Sense frame definition described hereinabove.

All future specifications are expected to include the this specificationas a Base Standard which all future specifications will support. TheRate Negotiation mechanism described provides for stations initiatingcommunication in the Base Standard and negotiating up to futurespecifications.

Those skilled in the art can appreciate that, while the presentinvention has been specifically described in conjunction with telephonelines in a home networking environment, other equivalent transmissionmedium could be used to implement the present invention. For example,the transmission medium for the frame-based communications network couldinclude power lines, or even wireless mediums, interconnectingtransmitting and receiving stations.

(Deference: Loop, looking for carrier sense, and when found determinewhether the transmission was a collision or valid frame. If it was acollision, process the signal slots and run the collision resolutionalgorithm. In any case, then process the priority slots, looking forcarrier. Note that the “current” priority level is sticky from the slotthe last collision occurred in. Note that the Backoff Level (BL) andMaximum Backoff Level (MBL) counters are saturating at 0 and 15.} ConstnPriorities = 8; {Number of priority levels} nSignals = 3; {Number ofsignal slots} nLevels = 16; {Number of Backoff Levels} processDeference; begin currentPriority := 0; {Priority of the slot we are in}cycle {deference loop} sawFrame := false; sawCollision := false; whilenot carrierSense( ) do nothing; {watch for carrier to appear} deferring:= true; startTime := time( ); stopTime := startTime; whilecarrierSense( ) do stopTime := time( ); if ((stopTime − startTime >CD_MIN) and (stopTime − startTime < CD_THRESHOLD)) or collisionSense( )then sawCollision := true else sawFrame := true; (After a collision,process the three signal slots} if sawCollision then begin {wait untilthe end of the IFG, timing from start of fragment reduces skew, sincestart-of-carrier uncertainty is less than end-of-carrier uncertainty}while (time( ) − startTime < CS_IFG + CD_FRAG) do nothing( );computeSignals( ); for (i := 0; i < nSignals; i++) begin startTime :=time( ); signal[i] := 0; if signalSlot = i then sendSignal( ); while(time( ) − startTime < SIG_SLOT) do if carrierSense( ) then signal[i] :=1; end; processSignals( ); end; if (not sawCollision) then begin {waituntil the end of the IFG} while (time( ) − stopTime < CS_IFG) donothing( ); {If last transmission was successful, drop Backoff Levels}BL[currentPriority] := saturate(0,nLevels− 1,BL[currentPriority]−1);MBL[currentPriority] := saturate(0,nlevels− 1,MBL[currentPriority]−1);end; {avoid timing hazard with transmitter, currentPriority must besetup before deferring is cleared} currentPriority := nPriorities−1;deferring := false; (Now time out the Priority (contention) slots} for(i := nPriorities−1; i>=0; i−−) begin slotTime := time( );currentPriority := i; while (time( ) −slotTime < PRI_SLOT) do ifcarrierSense( ) then endcycle; {restart deference loop} {if priorityslot passed with no contenders, then that priority level must be idle,good practice says make sure the backoff counters are reset}BL[currentPriority] := 0; MBL[currentPriority] := 0; end; end; {cycle}end; {Deference} {computeSignals: Determine which signals to send}function computeSignals( ); begin signalSlot := −1; {−1 means no signalto send, initialization} if (txReady and (txPriority = currentPriority)and BL[txPriority]=0) then signalSlot = integerRandom(nSignals); {selectBackoff Signal slot} end; {computeSignals} {processSignals: Process thereceived signals, adjusting the Backoff Levels} function processSignals(); begin psignals := 0; for (i=0; i < nSignals; i++) if signal[i] thenpsignals++; if (txReady and (txPriority = currentPriority)) then beginbackoffLevel := BL[currentPriority]; if backoffLevel = 0 then begin tem:= 0; for (i=0; i < signalSlot; i++) if signal[i] then tem++;BL[currentPriority] := saturate(0,nLevels−1,tem): end; if backoffLevel >0 then if psignals > 0 then BL[currentPriority] := saturate(0,nLevels−1,backoffLevel + psignals−1); end; if psignals > 0then begin if MBL[currentPriority] = 0 then MBL[currentPriority] :=psignals; else MBL[currentPriority] = saturate(0,nLevels−1,MBL[currentPriority] + psignals−1); end; end; {processSignals}{Transmitter: Wait for txReady and txPriority from the link levelprocess. send txFinished when frame has been sent.} process Transmitter;begin cycle while (not txReady) do nothing( ); BL[txPriority] :=MBL[txPriority]; while (not (txPriority >= currentPriority andBL[txPriority]=0) or deferring) do nothing( ); ttime := time( );xmtDataOn( ); {start data transmitting} while xmtBusy( ) and (time( ) −ttime < CD_FRAG) do begin if collisionSense( ) then begin xmtDataOff();{turn off, after sending minimum collision fragment} Ncollisions++;{timeout on excessive collision limit} if Ncollisions = attemptLimit−1then txFinished( ); endcycle; end; end; while xmtBusy( ) do nothing( );txReady := false; txFinished( ); {signal link level that frame has beentransmitted} end; { cycle } end; { Transmitter } {collisionSense: }function collisionSense( ); begin { When transmitting, detect thepresence of a second transmission. When receiving, detect overlappedtransmissions} end; { collisionSense } {Receiver: } process Receiver;begin { Wait for carrier sense. Demodulate received signals into frames.Reject collision fragments. Determine frame boundaries. Check FCS.Filter based on destination address. Perform optional Link Layersignaling and other controller functions.} end; { Receiver }

1. A signal embodied in a carrier wave for sending information fromtransmit stations to receive stations over a transmission medium of aframe-based based communications network, the information being sent intransmit frames having a frame format comprising a fixed rate header,followed by a variable rate payload, followed by a fixed rate trailer.2. The signal of claim 1, wherein the fixed rate header includes: apreamble; a frame control field; a destination address field; a sourceaddress field; and an ethertype field.
 3. The signal of claim 2, whereinthe preamble includes a repetition of four symbol sequences forfacilitating power estimation, gain control, baud frequency offsetestimation, equalizer training, carrier sensing and collision detection.4. The signal of claim 2, wherein the frame control field includesscrambler control information for frame scrambling initialization. 5.The signal of claim 2, wherein the frame control field includes apriority field to determine the absolute priority a transmit frame willhave when determining access to the transmission medium.
 6. The signalof claim 2, wherein the frame control field includes a payload encodingfield which determines constellation encoding of payload bits in thevariable rate payload.
 7. The signal of claim 2, wherein the framecontrol field includes a header check sequence for providing a cyclicredundancy check.
 8. The signal of claim 1, wherein the variable ratepayload is transmitted pursuant to dynamic adjustable frame encodingparameters for improving transmission performance for a transmit framebeing transmitted from a transmitting station to a receiving station. 9.A signal embodied in a carrier wave for sending information fromtransmit stations to receive stations over a transmission medium of aframe-based based communications network, the information being sent intransmit frames having a frame format comprising: a fixed rate header; avariable rate payload following the fixed rate header; and a fixed ratetrailer following the variable rate payload; wherein the fixed rateheader includes: a preamble, the preamble including a repetition of foursymbol sequences for facilitating power estimation, gain control, baudfrequency offset estimation, equalizer training, carrier sensing andcollision detection; a frame control field, the frame control fieldincluding: scrambler control information for frame scramblinginitialization, a priority field to determine the absolute priority atransmit frame will have when determining access to the transmissionmedium, a payload encoding field which determines constellation encodingof payload bits in the variable rate payload, and a header checksequence for providing a cyclic redundancy check; a destination addressfield; a source address field; and an ethertype field; and wherein thevariable rate payload is transmitted pursuant to dynamic adjustableframe encoding parameters for improving transmission performance for atransmit frame being transmitted from a transmitting station to areceiving station.
 10. A method for transmitting a transmitting frameembodied in a carrier wave from transmit stations to receive stationsover a transmission medium of a frame-based based communicationsnetwork, comprising: coupling one or more transmit stations to thetransmission medium, each transmit station transmitting frames having aframe format including a fixed rate header, followed by a variable ratepayload, followed by a fixed rate trailer; coupling one or more receivestations to the transmission medium, each receive station upon receivinga received frame corresponding to the transmitting frame addressed tothe receive station: detecting a start of the received frame utilizing apredefined preamble format for the transmitting frame having a pluralityof identical copies of a preamble symbol sequence transmittedsequentially; decoding the received frame; measuring and trackingperformance of frame decoding; determining network performancecharacteristics for establishing desired performance based uponmeasuring and tracking the performance of the frame decoding; indicatingto the transmit station changes to payload encoding parameters in thefixed rate header based upon determining network performance improvementcharacteristics, wherein the transmit station changes the payloadencoding parameters in the fixed rate header for encoding next futuretransmitting frames; and determining whether a collision between two ormore transmit stations occurred at one of the transmit stationsutilizing an estimate of error power of defined copies of the preamblesymbol sequence.
 11. The method of claim 10, wherein the fixed rateheader includes: a preamble, the preamble including a repetition of foursymbol sequences for facilitating power estimation, gain control, baudfrequency offset estimation, equalizer training, carrier sensing andcollision detection; a frame control field, the frame control fieldincluding: scrambler control information for frame scramblinginitialization, a priority field to determine the absolute priority atransmitting frame will have when determining access to the transmissionmedium, a payload encoding field which determines constellation encodingof payload bits in the variable rate payload, and a header checksequence for providing a cyclic redundancy check; a destination addressfield; a source address field; and an ethertype field; and wherein thevariable rate payload is transmitted pursuant to dynamic adjustableframe encoding parameters for improving transmission performance for atransmitting frame being transmitted from a transmitting station to areceiving station.